Broadcasting receiver and broadcast signal processing method

ABSTRACT

A digital broadcasting system which is robust against an error when mobile service data is transmitted and a method of processing data are disclosed. The mobile service data is subjected to an additional coding process and the coded mobile service data is transmitted. Accordingly, it is possible to cope with a serious channel variation while applying robustness to the mobile service data.

This application claims the benefit of the Korean Patent Application No.10-2008-0063683, filed on Jul. 1, 2008, which is hereby incorporated byreference as if fully set forth herein. Also, this application claimsthe benefit of U.S. Provisional Application No. 60/947,450, filed onJuly 2, 2007, which is hereby incorporated by reference. Thisapplication also claims the benefit of U.S. Provisional Application No.60/957,714, filed on Aug. 24, 2007, which is hereby incorporated byreference. This application also claims the benefit of U.S. ProvisionalApplication No. 60/974,084, filed on Sep. 21, 2007, which is herebyincorporated by reference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a digital broadcasting system, and moreparticularly, to a broadcasting receiver and a broadcast signalprocessing method.

2. Discussion of the Related Art

The Vestigial Sideband (VSB) transmission mode, which is adopted as thestandard for digital broadcasting in North America and the Republic ofKorea, is a system using a single carrier method. Therefore, thereceiving performance of the digital broadcast receiving system may bedeteriorated in a poor channel environment. Particularly, sinceresistance to changes in channels and noise is more highly required whenusing portable and/or mobile broadcast receivers, the receivingperformance may be even more deteriorated when transmitting mobileservice data by the VSB transmission mode.

SUMMARY OF THE INVENTION

Accordingly, the present invention is directed to a digital broadcastingsystem and a data processing method that substantially obviate one ormore problems due to limitations and disadvantages of the related art.

An object of the present invention is to provide a digital broadcastingsystem and a data processing method that are highly resistant to channelchanges and noise.

Another object of the present invention is to provide a digitalbroadcasting system and a data processing method that can enhance thereceiving performance of the receiving system by performing additionalencoding on mobile service data and by transmitting the processed datato the receiving system.

A further object of the present invention is to provide a digitalbroadcasting system and a data processing method that can also enhancethe receiving performance of the receiving system by inserting knowndata already known in accordance with a pre-agreement between thereceiving system and the transmitting system in a predetermined regionwithin a data region.

Additional advantages, objects, and features of the invention will beset forth in part in the description which follows and in part willbecome apparent to those having ordinary skill in the art uponexamination of the following or may be learned from practice of theinvention. The objectives and other advantages of the invention may berealized and attained by the structure particularly pointed out in thewritten description and claims hereof as well as the appended drawings.

To achieve these objects and other advantages and in accordance with thepurpose of the invention, as embodied and broadly described herein, adigital broadcast transmitting system may include a service multiplexerand a transmitter. The service multiplexer may multiplex mobile servicedata and main service data at a predetermined coding rate and maytransmit the multiplexed data to the transmitter. The transmitter mayperform additional encoding on the mobile service data being transmittedfrom the service multiplexer. The transmitter may also group a pluralityof additionally encoded mobile service data packets so as to form a datagroup. The transmitter may multiplex mobile service data packetsincluding mobile service data and main service data packets includingmain service data in packet units and may transmit the multiplexed datapackets to a digital broadcast receiving system.

Herein, the data group may be divided into a plurality of regionsdepending upon a degree of interference of the main service data. Also,a long known data sequence may be periodically inserted in regionswithout interference of the main service data. Also, a digital broadcastreceiving system according to an embodiment of the present invention maybe used for modulating and channel equalizing the known data sequence.

In another aspect of the present invention, a broadcasting receiverincludes a receiver which receives a broadcasting signal including mainservice data and mobile service data, wherein the mobile service dataconfigures a data group, and the data group is divided into a pluralityof regions, known data streams are linearly inserted into some of theplurality of regions, and initialization data used for initializing amemory in a trellis encoder of a transmitter is located at a startportion of the known data stream; a known data detector which detectsknown data in the received broadcasting signal; an equalizer whichchannel-equalizes the received mobile service data using the detectedknown data; and a decoder which extracts audio data and supplementaryinformation from the channel-equalized mobile service data, inverselyscales the audio data on the basis of a scale factor indicated by scalefactor index information included in the supplementary information, andrestores an audio signal. Herein N known data streams are inserted intosome of the plurality of regions and a transfer parameter is insertedbetween a first known data stream and a second known data stream amongthe N known data streams. And the mobile service data configures a RSframe, and the RS frame includes at least one data packet of the mobileservice data, a RS parity generated based on the at least one datapacket, and a CRC checksum generated based on the at least one datapacket and the RS parity. Herein the audio data is nonlinearlyquantized.

Also, the decoder performs a reciprocal exponentiation and inverselyquantizes the audio data. Herein the decoder performs the reciprocalexponentiation using 4/3 as a reciprocal exponent.

Also, the decoder inversely scales the audio data on the basis of thescale factor allocated to bands of the audio data.

Also, he decoder includes a bitstream formatter which extracts the audiodata and the supplementary information from the channel-equalized mobileservice data; a decoding and inverse quantization unit which inverselyscales the audio data on the basis of the scale factor indicated by thescale factor index information included in the supplementaryinformation; and an intensity/coupling unit which calculates stereoaudio data on the basis of information on a ratio of the power of theinversely scaled audio data to the power included in the supplementaryinformation.

In another aspect of the present invention, the method includesreceiving a broadcasting signal including main service data and mobileservice data; detecting known data in the received broadcasting signal;channel-equalizing the received mobile service data using the detectedknown data; and extracting audio data and supplementary information fromthe channel-equalized mobile service data, inversely scaling the audiodata on the basis of a scale factor indicated by scale factor indexinformation included in the supplementary information, and restoring anaudio signal, wherein the mobile service data configures a data group,and the data group is divided into a plurality of regions, known datastreams are linearly inserted into some of the plurality of regions, andinitialization data used for initializing a memory in a trellis encoderof a transmitter is located at a start portion of the known datastreams. Herein N known data streams are inserted into some of theplurality of regions and a transfer parameter is inserted between afirst known data stream and a second known data stream among the N knowndata streams. Herein the mobile service data configures a RS frame, andthe RS frame includes at least one data packet of the mobile servicedata, a RS parity generated based on the at least one data packet, and aCRC checksum generated based on the at least one data packet and the RSparity. Herein the audio data is nonlinearly quantized.

Also, the decoding step includes performing a reciprocal exponentiationand inversely quantizes the audio data.

Also, the decoding step includes inversely scaling the audio data on thebasis of the scale factor allocated to bands of the audio data.

It is to be understood that both the foregoing general description andthe following detailed description of the present invention areexemplary and explanatory and are intended to provide furtherexplanation of the invention as claimed.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings, which are included to provide a furtherunderstanding of the invention and are incorporated in and constitute apart of this application, illustrate embodiment(s) of the invention andtogether with the description serve to explain the principle of theinvention. In the drawings:

FIG. 1 illustrates a structure of a MPH frame for transmitting andreceiving mobile service data according to the present invention;

FIG. 2 illustrates an exemplary structure of a VSB frame;

FIG. 3 illustrates a mapping example of the positions to which the first4 slots of a sub-frame are assigned with respect to a VSB frame in aspace region;

FIG. 4 illustrates a mapping example of the positions to which the first4 slots of a sub-frame are assigned with respect to a VSB frame in atime region;

FIG. 5 illustrates an alignment of data after being data interleaved andidentified;

FIG. 6 illustrates an enlarged portion of the data group shown in FIG. 5for a better understanding of the present invention;

FIG. 7 illustrates an alignment of data before being data interleavedand identified;

FIG. 8 illustrates an enlarged portion of the data group shown in FIG. 7for a better understanding of the present invention;

FIG. 9 illustrates an exemplary assignement order of data groups beingassigned to one of 5 sub-frames according to the present invention;

FIG. 10 illustrates an example of multiple data groups of a singleparade being assigned (or allocated) to an MPH frame;

FIG. 11 illustrates an example of transmitting 3 parades to an MPH frameaccording to the present invention;

FIG. 12 illustrates an example of expanding the assignment process of 3parades to 5 sub-frames within an MPH frame;

FIG. 13 illustrates a block diagram showing a general structure of adigital broadcast transmitting system according to an embodiment of thepresent invention;

FIG. 14 illustrates a block diagram showing an example of a servicemultiplexer;

FIG. 15 illustrates a block diagram showing an example of a transmitteraccording to an embodiment of the present invention;

FIG. 16 illustrates a block diagram showing an example of apre-processor according to the present invention;

FIG. 17 illustrates a conceptual block diagram of the MPH frame encoderaccording to an embodiment of the present invention;

FIG. 18 illustrates a detailed block diagram of an RS frame encoderamong a plurality of RS frame encoders within an MPH frame encoder;

FIG. 19( a) and FIG. 19( b) illustrate a process of one or two RS framebeing divided into several portions, based upon an RS frame mode value,and a process of each portion being assigned to a corresponding regionwithin the respective data group;

FIG. 20( a) to FIG. 20( c) illustrate error correction encoding anderror detection encoding processes according to an embodiment of thepresent invention;

FIG. 21 illustrates an example of performing a row permutation (orinterleaving) process in super frame units according to the presentinvention;

FIG. 22( a) and FIG. 22( b) illustrate an example of creating an RSframe by grouping data, thereby performing error correction encoding anderror detection encoding;

FIG. 23( a) and FIG. 23( b) illustrate an exemplary process of dividingan RS frame for configuring a data group according to the presentinvention;

FIG. 24 illustrates a block diagram of a block processor according to anembodiment of the present invention;

FIG. 25 illustrates a detailed block diagram of a convolution encoder ofthe block processor of FIG. 24;

FIG. 26 illustrates a symbol interleaver of the block processor of FIG.24;

FIG. 27 illustrates a block diagram of a group formatter according to anembodiment of the present invention;

FIG. 28 illustrates a detailed diagram of one of 12 trellis encodersincluded in the trellis encoding module of FIG. 15;

FIG. 29 illustrates an example of assigning signaling information areaaccording to an embodiment of the present invention;

FIG. 30 illustrates a detailed block diagram of a signaling encoderaccording to the present invention;

FIG. 31 illustrates an example of a syntax structure of TPC dataaccording to the present invention;

FIG. 32 illustrates an example of power saving of in a receiver whentransmitting 3 parades to an MPH frame level according to the presentinvention;

FIG. 33 illustrates an example of a transmission scenario of the TPCdata and the FIC data level according to the present invention;

FIG. 34 illustrates an example of a training sequence at the byte levelaccording to the present invention;

FIG. 35 illustrates an example of a training sequence at the symbolaccording to the present invention;

FIG. 36 illustrates a block diagram of a demodulating unit in areceiving system according to the present invention;

FIG. 37 illustrates a data structure showing an example of known databeing periodically inserted in valid data according to the presentinvention;

FIG. 38 illustrates a block diagram showing a structure of a demodulatorof the demodulating unit shown in FIG. 36;

FIG. 39 illustrates a detailed block diagram of the demodulator shown inFIG. 38;

FIG. 40 illustrates a block diagram of a frequency offset estimatoraccording to an embodiment of the present invention;

FIG. 41 illustrates a block diagram of a known data detector and initialfrequency offset estimator according to the present invention;

FIG. 42 illustrates a block diagram of a partial correlator shown inFIG. 41;

FIG. 43 illustrates a second example of the timing recovery unitaccording to the present invention;

FIG. 44( a) and FIG. 44( b) illustrate examples of detecting timingerror in a time domain;

FIG. 45( a) and FIG. 45( b) illustrate other examples of detectingtiming error in a time domain;

FIG. 46 illustrates an example of detecting timing error usingcorrelation values of FIG. 44 and FIG. 45;

FIG. 47 illustrates an example of a timing error detector according tothe present invention;

FIG. 48 illustrates an example of detecting timing error in a frequencydomain according to an embodiment of the present invention;

FIG. 49 illustrates another example of a timing error detector accordingto the present invention;

FIG. 50 illustrates a block diagram of a DC remover according to anembodiment of the present invention;

FIG. 51 illustrates an example of shifting sample data inputted to a DCestimator shown in FIG. 50;

FIG. 52 illustrates a block diagram of a DC remover according to anotherembodiment of the present invention;

FIG. 53 illustrates a block diagram of another example of a channelequalizer according to the present invention;

FIG. 54 illustrates a detailed block diagram of an example of aremaining carrier phase error estimator according to the presentinvention;

FIG. 55 illustrates a block diagram of a phase error detector obtaininga remaining carrier phase error and phase noise according to the presentinvention;

FIG. 56 illustrates a phase compensator according to an embodiment ofthe present invention;

FIG. 57 illustrates a block diagram of another example of a channelequalizer according to the present invention;

FIG. 58 illustrates a block diagram of another example of a channelequalizer according to the present invention;

FIG. 59 illustrates a block diagram of another example of a channelequalizer according to the present invention;

FIG. 60 illustrates a block diagram of an example of a CIR estimatoraccording to the present invention;

FIG. 61 illustrates a block diagram of an example of a block decoderaccording to the present invention;

FIG. 62 illustrates a block diagram of an example of a feedbackdeformatter according to the present invention;

FIG. 63 to FIG. 65 illustrate process steps of error correction decodingaccording to an embodiment of the present invention;

FIG. 66 illustrates a block diagram of a receiving system according toan embodiment of the present invention;

FIG. 67 illustrates a bit stream syntax for a VCT according to thepresent invention;

FIG. 68 illustrates a service_type field according to an embodiment ofthe present invention;

FIG. 69 illustrates a service location descriptor according to anembodiment of the present invention;

FIG. 70 illustrates examples that may be assigned to the stream_typefield according to the present invention;

FIG. 71 illustrates a bit stream syntax for an EIT according to thepresent invention; and

FIG. 72 illustrates a block diagram of a receiving system according toanother embodiment of the present invention;

FIG. 73 is a block diagram showing an MPH receiver according to anembodiment of the present invention;

FIG. 74 is a view showing a method of compressing an audio signal and anaudio signal processing device for performing the method;

FIG. 75 is a view illustrating a masking effect used for compressing anaudio signal;

FIG. 76 is a block diagram showing the basic structure of a generalaudio encoder;

FIG. 77 is a view showing in detail an audio signal encoding apparatusof FIG. 76 according to an embodiment of the present invention;

FIG. 78 is a block diagram showing an audio signal decoding apparatus;

FIG. 79 is a view showing the basic configuration of the encodingapparatus according to the general MPEG standard;

FIG. 80 is a graph showing an SMR curve of sub-bands in a specific frameaccording to FIG. 79;

FIG. 81 is a block diagram showing an AAC encoding apparatus accordingto an embodiment of the present invention;

FIG. 82 is a block diagram showing an AAC decoding apparatus accordingto an embodiment of the present invention;

FIG. 83 is a flowchart illustrating a double structure for bitallocation and quantization according to an embodiment of the presentinvention;

FIG. 84 is a graph showing a bit allocation method according to anembodiment of the present invention;

FIG. 85 is a block diagram showing an MP3 encoding apparatus accordingto an embodiment of the present invention; and

FIG. 86( a) and FIG. 86( b) are a view showing equations forillustrating a joint stereo method according to an embodiment of thepresent invention.

FIG. 87 is a view showing another embodiment of the protocol stack forthe mobile service according to the present invention;

FIG. 88 is a view showing an example of an IP datagram generated by theprotocol stack shown in FIG. 87;

DETAILED DESCRIPTION OF THE INVENTION

Reference will now be made in detail to the preferred embodiments of thepresent invention, examples of which are illustrated in the accompanyingdrawings. Wherever possible, the same reference numbers will be usedthroughout the drawings to refer to the same or like parts. In addition,although the terms used in the present invention are selected fromgenerally known and used terms, some of the terms mentioned in thedescription of the present invention have been selected by the applicantat his or her discretion, the detailed meanings of which are describedin relevant parts of the description herein. Furthermore, it is requiredthat the present invention is understood, not simply by the actual termsused but by the meaning of each term lying within.

Among the terms used in the description of the present invention, mainservice data correspond to data that can be received by a fixedreceiving system and may include audio/video (A/V) data. Morespecifically, the main service data may include A/V data of highdefinition (HD) or standard definition (SD) levels and may also includediverse data types required for data broadcasting. Also, the known datacorrespond to data pre-known in accordance with a pre-arranged agreementbetween the receiving system and the transmitting system. Additionally,among the terms used in the present invention, “MPH” corresponds to theinitials of “mobile”, “pedestrian”, and “handheld” and represents theopposite concept of a fixed-type system. Furthermore, the MPH servicedata may include at least one of mobile service data, pedestrian servicedata, and handheld service data, and will also be referred to as “mobileservice data” for simplicity. Herein, the mobile service data not onlycorrespond to MPH service data but may also include any type of servicedata with mobile or portable characteristics.

Therefore, the mobile service data according to the present inventionare not limited only to the MPH service data.

The above-described mobile service data may correspond to data havinginformation, such as program execution files, stock information, and soon, and may also correspond to A/V data. Most particularly, the mobileservice data may correspond to A/V data having lower resolution andlower data rate as compared to the main service data. For example, if anA/V codec that is used for a conventional main service corresponds to aMPEG-2 codec, a MPEG-4 advanced video coding (AVC) or scalable videocoding (SVC) having better image compression efficiency may be used asthe A/V codec for the mobile service. Furthermore, any type of data maybe transmitted as the mobile service data. For example, transportprotocol expert group (TPEG) data for broadcasting real-timetransportation information may be transmitted as the main service data.

Also, a data service using the mobile service data may include weatherforecast services, traffic information services, stock informationservices, viewer participation quiz programs, real-time polls andsurveys, interactive education broadcast programs, gaming services,services providing information on synopsis, character, background music,and filming sites of soap operas or series, services providinginformation on past match scores and player profiles and achievements,and services providing information on product information and programsclassified by service, medium, time, and theme enabling purchase ordersto be processed. Herein, the present invention is not limited only tothe services mentioned above. In the present invention, the transmittingsystem provides backward compatibility in the main service data so as tobe received by the conventional receiving system. Herein, the mainservice data and the mobile service data are multiplexed to the samephysical channel and then transmitted.

Furthermore, the digital broadcast transmitting system according to thepresent invention performs additional encoding on the mobile servicedata and inserts the data already known by the receiving system andtransmitting system (e.g., known data), thereby transmitting theprocessed data. Therefore, when using the transmitting system accordingto the present invention, the receiving system may receive the mobileservice data during a mobile state and may also receive the mobileservice data with stability despite various distortion and noiseoccurring within the channel.

MPH Frame Structure

In the embodiment of the present invention, the mobile service data arefirst multiplexed with main service data in MPH frame units and, then,modulated in a VSB mode and transmitted to the receiving system. At thispoint, one MPH frame consists of K1 number of sub-frames, wherein onesub-frame includes K2 number of slots. Also, each slot may be configuredof K3 number of data packets. In the embodiment of the presentinvention, K1 will be set to 5, K2 will be set to 16, and K3 will be setto 156 (i.e., K1=5, K2=16, and K3=156). The values for K1, K2, and K3presented in this embodiment either correspond to values according to apreferred embodiment or are merely exemplary. Therefore, theabove-mentioned values will not limit the scope of the presentinvention.

FIG. 1 illustrates a structure of a MPH frame for transmitting andreceiving mobile service data according to the present invention. In theexample shown in FIG. 1, one MPH frame consists of 5 sub-frames, whereineach sub-frame includes 16 slots. In this case, the MPH frame accordingto the present invention includes 5 sub-frames and 80 slots. Also, in apacket level, one slot is configured of 156 data packets (i.e.,transport stream packets), and in a symbol level, one slot is configuredof 156 data segments. Herein, the size of one slot corresponds to onehalf (½) of a VSB field. More specifically, since one 207-byte datapacket has the same amount of data as a data segment, a data packetprior to being interleaved may also be used as a data segment. At thispoint, two VSB fields are grouped to form a VSB frame.

FIG. 2 illustrates an exemplary structure of a VSB frame, wherein oneVSB frame consists of 2 VSB fields (i.e., an odd field and an evenfield). Herein, each VSB field includes a field synchronization segmentand 312 data segments. The slot corresponds to a basic time period formultiplexing the mobile service data and the main service data. Herein,one slot may either include the mobile service data or be configuredonly of the main service data. If one MPH frame is transmitted duringone slot, the first 118 data packets within the slot correspond to adata group. And, the remaining 38 data packets become the main servicedata packets. In another example, when no data group exists in a slot,the corresponding slot is configured of 156 main service data packets.Meanwhile, when the slots are assigned to a VSB frame, an off-set existsfor each assigned position.

FIG. 3 illustrates a mapping example of the positions to which the first4 slots of a sub-frame are assigned with respect to a VSB frame in aspace region. And, FIG. 4 illustrates a mapping example of the positionsto which the first 4 slots of a sub-frame are assigned with respect to aVSB frame in a time region. Referring to FIG. 3 and FIG. 4, a 38^(th)data packet (TS packet #37) of a 1^(st) slot (Slot #0) is mapped to the1^(st) data packet of an odd VSB field. A 38^(th) data packet (TS packet#37) of a 2^(nd) slot (Slot #1) is mapped to the 157^(th) data packet ofan odd VSB field. Also, a 38^(th) data packet (TS packet #37) of a3^(rd) slot (Slot #2) is mapped to the 1^(st) data packet of an even VSBfield. And, a 38^(th) data packet (TS packet #37) of a 4^(th) slot (Slot#3) is mapped to the 157^(th) data packet of an even VSB field.Similarly, the remaining 12 slots within the corresponding sub-frame aremapped in the subsequent VSB frames using the same method.

Meanwhile, one data group may be divided into at least one or morehierarchical regions. And, depending upon the characteristics of eachhierarchical region, the type of mobile service data being inserted ineach region may vary. For example, the data group within each region maybe divided (or categorized) based upon the receiving performance. In anexample given in the present invention, a data group is divided intoregions A, B, C, and D in a data configuration prior to datadeinterleaving.

FIG. 5 illustrates an alignment of data after being data interleaved andidentified. FIG. 6 illustrates an enlarged portion of the data groupshown in FIG. 5 for a better understanding of the present invention.FIG. 7 illustrates an alignment of data before being data interleavedand identified. And, FIG. 8 illustrates an enlarged portion of the datagroup shown in FIG. 7 for a better understanding of the presentinvention. More specifically, a data structure identical to that shownin FIG. 5 is transmitted to a receiving system. In other words, one datapacket is data-interleaved so as to be scattered to a plurality of datasegments, thereby being transmitted to the receiving system. FIG. 5illustrates an example of one data group being scattered to 170 datasegments. At this point, since one 207-byte packet has the same amountof data as one data segment, the packet that is not yet processed withdata-interleaving may be used as the data segment.

FIG. 5 shows an example of dividing a data group prior to beingdata-interleaved into 10 MPH blocks (i.e., MPH block 1 (B1) to MPH block10 (B10)). In this example, each MPH block has the length of 16segments. Referring to FIG. 5, only the RS parity data are allocated toportions of the first 5 segments of the MPH block 1 (B1) and the last 5segments of the MPH block 10 (B10). The RS parity data are excluded inregions A to D of the data group. More specifically, when it is assumedthat one data group is divided into regions A, B, C, and D, each MPHblock may be included in any one of region A to region D depending uponthe characteristic of each MPH block within the data group.

Herein, the data group is divided into a plurality of regions to be usedfor different purposes. More specifically, a region of the main servicedata having no interference or a very low interference level may beconsidered to have a more resistant (or stronger) receiving performanceas compared to regions having higher interference levels. Additionally,when using a system inserting and transmitting known data in the datagroup, wherein the known data are known based upon an agreement betweenthe transmitting system and the receiving system, and when consecutivelylong known data are to be periodically inserted in the mobile servicedata, the known data having a predetermined length may be periodicallyinserted in the region having no interference from the main service data(i.e., a region wherein the main service data are not mixed). However,due to interference from the main service data, it is difficult toperiodically insert known data and also to insert consecutively longknown data to a region having interference from the main service data.

Referring to FIG. 5, MPH block 4 (B4) to MPH block 7 (B7) correspond toregions without interference of the main service data. MPH block 4 (B4)to MPH block 7 (B7) within the data group shown in FIG. 5 correspond toa region where no interference from the main service data occurs. Inthis example, a long known data sequence is inserted at both thebeginning and end of each MPH block. In the description of the presentinvention, the region including MPH block 4 (B4) to MPH block 7 (B7)will be referred to as “region A (=B4+B5+B6+B7)”. As described above,when the data group includes region A having a long known data sequenceinserted at both the beginning and end of each MPH block, the receivingsystem is capable of performing equalization by using the channelinformation that can be obtained from the known data. Therefore, thestrongest equalizing performance may be yielded (or obtained) from oneof region A to region D.

In the example of the data group shown in FIG. 5, MPH block 3 (B3) andMPH block 8 (B8) correspond to a region having little interference fromthe main service data. Herein, a long known data sequence is inserted inonly one side of each MPH block B3 and B8. More specifically, due to theinterference from the main service data, a long known data sequence isinserted at the end of MPH block 3 (B3), and another long known datasequence is inserted at the beginning of MPH block 8 (B8). In thepresent invention, the region including MPH block 3 (B3) and MPH block 8(B8) will be referred to as “region B(=B3+B8)”. As described above, whenthe data group includes region B having a long known data sequenceinserted at only one side (beginning or end) of each MPH block, thereceiving system is capable of performing equalization by using thechannel information that can be obtained from the known data. Therefore,a stronger equalizing performance as compared to region C/D may beyielded (or obtained).

Referring to FIG. 5, MPH block 2 (B2) and MPH block 9 (B9) correspond toa region having more interference from the main service data as comparedto region B. A long known data sequence cannot be inserted in any sideof MPH block 2 (B2) and MPH block 9 (B9). Herein, the region includingMPH block 2 (B2) and MPH block 9 (B9) will be referred to as “regionC(=B2+B9)”. Finally, in the example shown in FIG. 5, MPH block 1 (B1)and MPH block 10 (B10) correspond to a region having more interferencefrom the main service data as compared to region C. Similarly, a longknown data sequence cannot be inserted in any side of MPH block 1 (B1)and MPH block 10 (B10). Herein, the region including MPH block 1 (B1)and MPH block 10 (B10) will be referred to as “region D (=B1+B10)”.Since region C/D is spaced further apart from the known data sequence,when the channel environment undergoes frequent and abrupt changes, thereceiving performance of region C/D may be deteriorated.

FIG. 7 illustrates a data structure prior to data interleaving. Morespecifically, FIG. 7 illustrates an example of 118 data packets beingallocated to a data group. FIG. 7 shows an example of a data groupconsisting of 118 data packets, wherein, based upon a reference packet(e.g., a 1^(st) packet (or data segment) or 157^(th) packet (or datasegment) after a field synchronization signal), when allocating datapackets to a VSB frame, 37 packets are included before the referencepacket and 81 packets (including the reference packet) are includedafterwards. In other words, with reference to FIG. 5, a fieldsynchronization signal is placed (or assigned) between MPH block 2 (B2)and MPH block 3 (B3). Accordingly, this indicates that the slot has anoff-set of 37 data packets with respect to the corresponding VSB field.The size of the data groups, number of hierarchical regions within thedata group, the size of each region, the number of MPH blocks includedin each region, the size of each MPH block, and so on described aboveare merely exemplary. Therefore, the present invention will not belimited to the examples described above.

FIG. 9 illustrates an exemplary assignement order of data groups beingassigned to one of 5 sub-frames, wherein the 5 sub-frames configure anMPH frame. For example, the method of assigning data groups may beidentically applied to all MPH frames or differently applied to each MPHframe. Furthermore, the method of assinging data groups may beidentically applied to all sub-frames or differently applied to eachsub-frame. At this point, when it is assumed that the data groups areassigned using the same method in all sub-frames of the correspondingMPH frame, the total number of data groups being assigned to an MPHframe is equal to a multiple of ‘5’. According to the embodiment of thepresent invention, a plurality of consecutive data groups is assigned tobe spaced as far apart from one another as possible within the MPHframe. Thus, the system can be capable of responding promptly andeffectively to any burst error that may occur within a sub-frame.

For example, when it is assumed that 3 data groups are assigned to asub-frame, the data groups are assigned to a 1^(st) slot (Slot #0), a5^(th) slot (Slot #4), and a 9^(th) slot (Slot #8) in the sub-frame,respectively. FIG. 9 illustrates an example of assigning 16 data groupsin one sub-frame using the above-described pattern (or rule). In otherwords, each data group is serially assigned to 16 slots corresponding tothe following numbers: 0, 8, 4, 12, 1, 9, 5, 13, 2, 10, 6, 14, 3, 11, 7,and 15. Equation 1 below shows the above-described rule (or pattern) forassigning data groups in a sub-frame.

$\begin{matrix}\begin{matrix}{j = \left( {{4i} + 0} \right)} & {mod} & 16 & \; & \; \\\; & {0 = 0} & {if} & {{i < 4},} & \; \\\; & {0 = 2} & {else} & {if} & {{i < 8},} \\{{Herein},} & {0 = 1} & {else} & {if} & {{i < 12},} \\\; & {0 = 3} & {{else}.} & \; & \;\end{matrix} & {{Equation}\mspace{14mu} 1}\end{matrix}$

Herein, j indicates the slot number within a sub-frame. The value of jmay range from 0 to 15 (i.e., 0≦j≦15). Also, variable i indicates thedata group number. The value of i may range from 0 to 15 (i.e., 0≦i≦15).

In the present invention, a collection of data groups included in a MPHframe will be referred to as a “parade”. Based upon the RS frame mode,the parade transmits data of at least one specific RS frame. The mobileservice data within one RS frame may be assigned either to all ofregions A/B/C/D within the corresponding data group, or to at least oneof regions A/B/C/D. In the embodiment of the present invention, themobile service data within one RS frame may be assigned either to all ofregions A/B/C/D, or to at least one of regions A/B and regions C/D. Ifthe mobile service data are assigned to the latter case (i.e., one ofregions A/B and regions C/D), the RS frame being assigned to regions A/Band the RS frame being assigned to regions C/D within the correspondingdata group are different from one another.

In the description of the present invention, the RS frame being assignedto regions A/B within the corresponding data group will be referred toas a “primary RS frame”, and the RS frame being assigned to regions C/Dwithin the corresponding data group will be referred to as a “secondaryRS frame”, for simplicity. Also, the primary RS frame and the secondaryRS frame form (or configure) one parade. More specifically, when themobile service data within one RS frame are assigned either to all ofregions A/B/C/D within the corresponding data group, one paradetransmits one RS frame. Conversely, when the mobile service data withinone RS frame are assigned either to at least one of regions A/B andregions C/D, one parade may transmit up to 2 RS frames. Morespecifically, the RS frame mode indicates whether a parade transmits oneRS frame, or whether the parade transmits two RS frames. Table 1 belowshows an example of the RS frame mode.

TABLE 1 RS frame mode (2 bits) Description 00 There is only one primaryRS frame for all group regions 01 There are two separate RS frames.Primary RS frame for group regions A and B Secondary RS frame for groupregions C and D 10 Reserved 11 Reserved

Table 1 illustrates an example of allocating 2 bits in order to indicatethe RS frame mode. For example, referring to Table 1, when the RS framemode value is equal to ‘00’, this indicates that one parade transmitsone RS frame. And, when the RS frame mode value is equal to ‘01’, thisindicates that one parade transmits two RS frames, i.e., the primary RSframe and the secondary RS frame. More specifically, when the RS framemode value is equal to ‘01’, data of the primary RS frame for regionsA/B are assigned and transmitted to regions A/B of the correspondingdata group. Similarly, data of the secondary RS frame for regions C/Dare assigned and transmitted to regions C/D of the corresponding datagroup.

Additionally, one RS frame transmits one ensemble. Herein, the ensembleis a collection of services requiring the same quality of service (QOS)and being encoded with the same FEC codes. More specifically, when oneparade is configured of one RS frame, then one parade transmits oneensemble. Conversely, when one parade is configured of two RS frames,i.e., when one parade is configured of a primary RS frame and asecondary RS frame, then one parade transmits two ensembles (i.e., aprimary ensemble and a secondary ensemble). More specifically, theprimary ensemble is transmitted through a primary RS frame of a parade,and the secondary ensemble is transmitted through a secondary RS frameof a parade. The RS frame is a 2-dimensional data frame through which anensemble is RS-CRC encoded.

As described in the assignment of data groups, the parades are alsoassigned to be spaced as far apart from one another as possible withinthe sub-frame. Thus, the system can be capable of responding promptlyand effectively to any burst error that may occur within a sub-frame.Furthermore, the method of assinging parades may be identically appliedto all sub-frames or differently applied to each sub-frame. According tothe embodiment of the present invention, the parades may be assigneddifferently for each MPH frame and identically for all sub-frames withinan MPH frame. More specifically, the MPH frame structure may vary by MPHframe units. Thus, an ensemble rate may be adjusted on a more frequentand flexible basis.

FIG. 10 illustrates an example of multiple data groups of a singleparade being assigned (or allocated) to an MPH frame. More specifically,FIG. 10 illustrates an example of a plurality of data groups included ina single parade, wherein the number of data groups included in asub-frame is equal to ‘3’, being allocated to an MPH frame. Referring toFIG. 10, 3 data groups are sequentially assigned to a sub-frame at acycle period of 4 slots. Accordingly, when this process is equallyperformed in the 5 sub-frames included in the corresponding MPH frame,15 data groups are assigned to a single MPH frame. Herein, the 15 datagroups correspond to data groups included in a parade. Therefore, sinceone sub-frame is configured of 4 VSB frame, and since 3 data groups areincluded in a sub-frame, the data group of the corresponding parade isnot assigned to one of the 4 VSB frames within a sub-frame.

For example, when it is assumed that one parade transmits one RS frame,and that a RS frame encoder located in a later block performsRS-encoding on the corresponding RS frame, thereby adding 24 bytes ofparity data to the corresponding RS frame and transmitting the processedRS frame, the parity data occupy approximately 11.37% (=24/(187+24)×100)of the total code word length. Meanwhile, when one sub-frame includes 3data groups, and when the data groups included in the parade areassigned, as shown in FIG. 10, a total of 15 data groups form an RSframe. Accordingly, even when an error occurs in an entire data groupdue to a burst noise within a channel, the percentile is merely 6.67%(=1/15×100). Therefore, the receiving system may correct all errors byperforming an erasure RS decoding process. More specifically, when theerasure RS decoding is performed, a number of channel errorscorresponding to the number of RS parity bytes may be corrected. Bydoing so, the receiving system may correct the error of at least onedata group within one parade. Thus, the minimum burst noise lengthcorrectable by a RS frame is over 1 VSB frame.

Meanwhile, when data groups of a parade are assigned as described above,either main service data may be assigned between each data group, ordata groups corresponding to different parades may be assigned betweeneach data group. More specifically, data groups corresponding tomultiple parades may be assigned to one MPH frame. Basically, the methodof assigning data groups corresponding to multiple parades is verysimilar to the method of assigning data groups corresponding to a singleparade. In other words, data groups included in other parades that areto be assigned to an MPH frame are also respectively assigned accordingto a cycle period of 4 slots. At this point, data groups of a differentparade may be sequentially assigned to the respective slots in acircular method. Herein, the data groups are assigned to slots startingfrom the ones to which data groups of the previous parade have not yetbeen assigned. For example, when it is assumed that data groupscorresponding to a parade are assigned as shown in FIG. 10, data groupscorresponding to the next parade may be assigned to a sub-frame startingeither from the 12^(th) slot of a sub-frame. However, this is merelyexemplary. In another example, the data groups of the next parade mayalso be sequentially assigned to a different slot within a sub-frame ata cycle period of 4 slots starting from the 3^(rd) slot.

FIG. 11 illustrates an example of transmitting 3 parades (Parade #0,Parade #1, and Parade #2) to an MPH frame. More specifically, FIG. 11illustrates an example of transmitting parades included in one of 5sub-frames, wherein the 5 sub-frames configure one MPH frame. When the1^(st) parade (Parade #0) includes 3 data groups for each sub-frame, thepositions of each data groups within the sub-frames may be obtained bysubstituting values ‘0’ to ‘2’ for i in Equation 1. More specifically,the data groups of the 1^(st) parade (Parade #0) are sequentiallyassigned to the 1^(st), 5^(th), and 9^(th) slots (Slot #0, Slot #4, andSlot #8) within the sub-frame. Also, when the 2^(nd) parade includes 2data groups for each sub-frame, the positions of each data groups withinthe sub-frames may be obtained by substituting values ‘3’ and ‘4’ for iin Equation 1. More specifically, the data groups of the 2^(nd) parade(Parade #1) are sequentially assigned to the 2^(nd) and 12^(th) slots(Slot #3 and Slot #11) within the sub-frame. Finally, when the 3^(rd)parade includes 2 data groups for each sub-frame, the positions of eachdata groups within the sub-frames may be obtained by substituting values‘5’ and ‘6’ for i in Equation 1. More specifically, the data groups ofthe 3^(rd) parade (Parade #2) are sequentially assigned to the 7^(th)and 11^(th) slots (Slot #6 and Slot #10) within the sub-frame.

As described above, data groups of multiple parades may be assigned to asingle MPH frame, and, in each sub-frame, the data groups are seriallyallocated to a group space having 4 slots from left to right. Therefore,a number of groups of one parade per sub-frame (NOG) may correspond toany one integer from ‘1’ to ‘8’. Herein, since one MPH frame includes 5sub-frames, the total number of data groups within a parade that can beallocated to an MPH frame may correspond to any one multiple of ‘5’ranging from ‘5’ to ‘40’.

FIG. 12 illustrates an example of expanding the assignment process of 3parades, shown in FIG. 11, to 5 sub-frames within an MPH frame.

General Description of The Transmitting System

FIG. 13 illustrates a block diagram showing a general structure of adigital broadcast transmitting system according to an embodiment of thepresent invention.

Herein, the digital broadcast transmitting includes a servicemultiplexer 100 and a transmitter 200. Herein, the service multiplexer100 is located in the studio of each broadcast station, and thetransmitter 200 is located in a site placed at a predetermined distancefrom the studio. The transmitter 200 may be located in a plurality ofdifferent locations. Also, for example, the plurality of transmittersmay share the same frequency. And, in this case, the plurality oftransmitters receives the same signal. Accordingly, in the receivingsystem, a channel equalizer may compensate signal distortion, which iscaused by a reflected wave, so as to recover the original signal. Inanother example, the plurality of transmitters may have differentfrequencies with respect to the same channel.

A variety of methods may be used for data communication each of thetransmitters, which are located in remote positions, and the servicemultiplexer. For example, an interface standard such as a synchronousserial interface for transport of MPEG-2 data (SMPTE-310M). In theSMPTE-310M interface standard, a constant data rate is decided as anoutput data rate of the service multiplexer. For example, in case of the8VSB mode, the output data rate is 19.39 Mbps, and, in case of the 16VSBmode, the output data rate is 38.78 Mbps. Furthermore, in theconventional 8VSB mode transmitting system, a transport stream (TS)packet having a data rate of approximately 19.39 Mbps may be transmittedthrough a single physical channel. Also, in the transmitting systemaccording to the present invention provided with backward compatibilitywith the conventional transmitting system, additional encoding isperformed on the mobile service data. Thereafter, the additionallyencoded mobile service data are multiplexed with the main service datato a TS packet form, which is then transmitted. At this point, the datarate of the multiplexed TS packet is approximately 19.39 Mbps.

At this point, the service multiplexer 100 receives at least one type ofmobile service data and program specific information/program and systeminformation protocol (PSI/PSIP) table data for each mobile service so asto encapsulate the received data to each TS packet. Also, the servicemultiplexer 100 receives at least one type of main service data andPSI/PSIP table data for each main service and encapsulates the receiveddata to a transport stream (TS) packet. Subsequently, the TS packets aremultiplexed according to a predetermined multiplexing rule and outputsthe multiplexed packets to the transmitter 200.

Service Multiplexer

FIG. 14 illustrates a block diagram showing an example of the servicemultiplexer. The service multiplexer includes a controller 110 forcontrolling the overall operations of the service multiplexer, aPSI/PSIP generator 120 for the main service, a PSI/PSIP generator 130for the mobile service, a null packet generator 140, a mobile servicemultiplexer 150, and a transport multiplexer 160.

The transport multiplexer 160 may include a main service multiplexer 161and a transport stream (TS) packet multiplexer 162.

Referring to FIG. 14, at least one type of compression encoded mainservice data and the PSI/PSIP table data generated from the PSI/PSIPgenerator 120 for the main service are inputted to the main servicemultiplexer 161 of the transport multiplexer 160. The main servicemultiplexer 161 encapsulates each of the inputted main service data andPSI/PSIP table data to MPEG-2 TS packet forms. Then, the MPEG-2 TSpackets are multiplexed and outputted to the TS packet multiplexer 162.Herein, the data packet being outputted from the main servicemultiplexer 161 will be referred to as a main service data packet forsimplicity.

Thereafter, at least one type of the compression encoded mobile servicedata and the PSI/PSIP table data generated from the PSI/PSIP generator130 for the mobile service are inputted to the mobile servicemultiplexer 150.

The mobile service multiplexer 150 encapsulates each of the inputtedmobile service data and PSI/PSIP table data to MPEG-2 TS packet forms.Then, the MPEG-2 TS packets are multiplexed and outputted to the TSpacket multiplexer 162. Herein, the data packet being outputted from themobile service multiplexer 150 will be referred to as a mobile servicedata packet for simplicity.

At this point, the transmitter 200 requires identification informationin order to identify and process the main service data packet and themobile service data packet. Herein, the identification information mayuse values pre-decided in accordance with an agreement between thetransmitting system and the receiving system, or may be configured of aseparate set of data, or may modify predetermined location value with inthe corresponding data packet.

As an example of the present invention, a different packet identifier(PID) may be assigned to identify each of the main service data packetand the mobile service data packet.

In another example, by modifying a synchronization data byte within aheader of the mobile service data, the service data packet may beidentified by using the synchronization data byte value of thecorresponding service data packet. For example, the synchronization byteof the main service data packet directly outputs the value decided bythe ISO/IEC13818-1 standard (i.e., 0x47) without any modification. Thesynchronization byte of the mobile service data packet modifies andoutputs the value, thereby identifying the main service data packet andthe mobile service data packet. Conversely, the synchronization byte ofthe main service data packet is modified and outputted, whereas thesynchronization byte of the mobile service data packet is directlyoutputted without being modified, thereby enabling the main service datapacket and the mobile service data packet to be identified.

A plurality of methods may be applied in the method of modifying thesynchronization byte. For example, each bit of the synchronization bytemay be inversed, or only a portion of the synchronization byte may beinversed.

As described above, any type of identification information may be usedto identify the main service data packet and the mobile service datapacket. Therefore, the scope of the present invention is not limitedonly to the example set forth in the description of the presentinvention.

Meanwhile, a transport multiplexer used in the conventional digitalbroadcasting system may be used as the transport multiplexer 160according to the present invention. More specifically, in order tomultiplex the mobile service data and the main service data and totransmit the multiplexed data, the data rate of the main service islimited to a data rate of (19.39-K) Mbps. Then, K Mbps, whichcorresponds to the remaining data rate, is assigned as the data rate ofthe mobile service. Thus, the transport multiplexer which is alreadybeing used may be used as it is without any modification.

Herein, the transport multiplexer 160 multiplexes the main service datapacket being outputted from the main service multiplexer 161 and themobile service data packet being outputted from the mobile servicemultiplexer 150. Thereafter, the transport multiplexer 160 transmits themultiplexed data packets to the transmitter 200.

However, in some cases, the output data rate of the mobile servicemultiplexer 150 may not be equal to K Mbps. In this case, the mobileservice multiplexer 150 multiplexes and outputs null data packetsgenerated from the null packet generator 140 so that the output datarate can reach K Mbps. More specifically, in order to match the outputdata rate of the mobile service multiplexer 150 to a constant data rate,the null packet generator 140 generates null data packets, which arethen outputted to the mobile service multiplexer 150.

For example, when the service multiplexer 100 assigns K Mbps of the19.39 Mbps to the mobile service data, and when the remaining (19.39-K)Mbps is, therefore, assigned to the main service data, the data rate ofthe mobile service data that are multiplexed by the service multiplexer100 actually becomes lower than K Mbps. This is because, in case of themobile service data, the pre-processor of the transmitting systemperforms additional encoding, thereby increasing the amount of data.Eventually, the data rate of the mobile service data, which may betransmitted from the service multiplexer 100, becomes smaller than KMbps.

For example, since the pre-processor of the transmitter performs anencoding process on the mobile service data at a coding rate of at least½, the amount of the data outputted from the pre-processor is increasedto more than twice the amount of the data initially inputted to thepre-processor. Therefore, the sum of the data rate of the main servicedata and the data rate of the mobile service data, both beingmultiplexed by the service multiplexer 100, becomes either equal to orsmaller than 19.39 Mbps.

Therefore, in order to match the data rate of the data that are finallyoutputted from the service multiplexer 100 to a constant data rate(e.g., 19.39 Mbps), an amount of null data packets corresponding to theamount of lacking data rate is generated from the null packet generator140 and outputted to the mobile service multiplexer 150.

Accordingly, the mobile service multiplexer 150 encapsulates each of themobile service data and the PSI/PSIP table data that are being inputtedto a MPEG-2 TS packet form. Then, the above-described TS packets aremultiplexed with the null data packets and, then, outputted to the TSpacket multiplexer 162.

Thereafter, the TS packet multiplexer 162 multiplexes the main servicedata packet being outputted from the main service multiplexer 161 andthe mobile service data packet being outputted from the mobile servicemultiplexer 150 and transmits the multiplexed data packets to thetransmitter 200 at a data rate of 19.39 Mbps.

According to an embodiment of the present invention, the mobile servicemultiplexer 150 receives the null data packets. However, this is merelyexemplary and does not limit the scope of the present invention. Inother words, according to another embodiment of the present invention,the TS packet multiplexer 162 may receive the null data packets, so asto match the data rate of the finally outputted data to a constant datarate. Herein, the output path and multiplexing rule of the null datapacket is controlled by the controller 110. The controller 110 controlsthe multiplexing processed performed by the mobile service multiplexer150, the main service multiplexer 161 of the transport multiplexer 160,and the TS packet multiplexer 162, and also controls the null datapacket generation of the null packet generator 140. At this point, thetransmitter 200 discards the null data packets transmitted from theservice multiplexer 100 instead of transmitting the null data packets.

Further, in order to allow the transmitter 200 to discard the null datapackets transmitted from the service multiplexer 100 instead oftransmitting them, identification information for identifying the nulldata packet is required. Herein, the identification information may usevalues pre-decided in accordance with an agreement between thetransmitting system and the receiving system. For example, the value ofthe synchronization byte within the header of the null data packet maybe modified so as to be used as the identification information.Alternatively, a transport_error_indicator flag may also be used as theidentification information.

In the description of the present invention, an example of using thetransport_error_indicator flag as the identification information will begiven to describe an embodiment of the present invention. In this case,the transport_error_indicator flag of the null data packet is set to‘1’, and the transport_error_indicator flag of the remaining datapackets are reset to ‘0’, so as to identify the null data packet. Morespecifically, when the null packet generator 140 generates the null datapackets, if the transport_error_indicator flag from the header field ofthe null data packet is set to ‘1’ and then transmitted, the null datapacket may be identified and, therefore, be discarded. In the presentinvention, any type of identification information for identifying thenull data packets may be used. Therefore, the scope of the presentinvention is not limited only to the examples set forth in thedescription of the present invention.

According to another embodiment of the present invention, a transmissionparameter may be included in at least a portion of the null data packet,or at least one table or an operations and maintenance (OM) packet (orOMP) of the PSI/PSIP table for the mobile service. In this case, thetransmitter 200 extracts the transmission parameter and outputs theextracted transmission parameter to the corresponding block and alsotransmits the extracted parameter to the receiving system if required.More specifically, a packet referred to as an OMP is defined for thepurpose of operating and managing the transmitting system. For example,the OMP is configured in accordance with the MPEG-2 TS packet format,and the corresponding PID is given the value of 0x1FFA. The OMP isconfigured of a 4-byte header and a 184-byte payload. Herein, among the184 bytes, the first byte corresponds to an OM type field, whichindicates the type of the OM packet.

In the present invention, the transmission parameter may be transmittedin the form of an OMP. And, in this case, among the values of thereserved fields within the OM_type field, a pre-arranged value is used,thereby indicating that the transmission parameter is being transmittedto the transmitter 200 in the form of an OMP. More specifically, thetransmitter 200 may find (or identify) the OMP by referring to the PID.Also, by parsing the OM_type field within the OMP, the transmitter 200can verify whether a transmission parameter is included after theOM_type field of the corresponding packet. The transmission parametercorresponds to supplemental data required for processing mobile servicedata from the transmitting system and the receiving system.

The transmission parameter corresponds to supplemental data required forprocessing mobile service data from the transmitting system and thereceiving system. Herein, the transmission parameter may include datagroup information, region information within the data group, blockinformation, RS frame information, super frame information, MPH frameinformation, parade information, ensemble information, informationassociated with serial concatenated convolution code (SCCC), and RS codeinformation. The significance of some information within thetransmission parameters has already been described in detail.Descriptions of other information that have not yet been described willbe in detail in a later process.

The transmission parameter may also include information on how signalsof a symbol domain are encoded in order to transmit the mobile servicedata, and multiplexing information on how the main service data and themobile service data or various types of mobile service data aremultiplexed.

The information included in the transmission parameter are merelyexemplary to facilitate the understanding of the present invention. And,the adding and deleting of the information included in the transmissionparameter may be easily modified and changed by anyone skilled in theart. Therefore, the present invention is not limited to the examplesproposed in the description set forth herein.

Furthermore, the transmission parameters may be provided from theservice multiplexer 100 to the transmitter 200. Alternatively, thetransmission parameters may also be set up by an internal controller(not shown) within the transmitter 200 or received from an externalsource.

Transmitter

FIG. 15 illustrates a block diagram showing an example of thetransmitter 200 according to an embodiment of the present invention.Herein, the transmitter 200 includes a controller 200, a demultiplexer210, a packet jitter mitigator 220, a pre-processor 230, a packetmultiplexer 240, a post-processor 250, a synchronization (sync)multiplexer 260, and a transmission unit 270. Herein, when a data packetis received from the service multiplexer 100, the demultiplexer 210should identify whether the received data packet corresponds to a mainservice data packet, a mobile service data packet, or a null datapacket. For example, the demultiplexer 210 uses the PID within thereceived data packet so as to identify the main service data packet andthe mobile service data packet. Then, the demultiplexer 210 uses atransport_error_indicator field to identify the null data packet. Themain service data packet identified by the demultiplexer 210 isoutputted to the packet jitter mitigator 220, the mobile service datapacket is outputted to the pre-processor 230, and the null data packetis discarded. If a transmission parameter is included in the null datapacket, then the transmission parameter is first extracted and outputtedto the corresponding block. Thereafter, the null data packet isdiscarded.

The pre-processor 230 performs an additional encoding process of themobile service data included in the service data packet, which isdemultiplexed and outputted from the demultiplexer 210. Thepre-processor 230 also performs a process of configuring a data group sothat the data group may be positioned at a specific place in accordancewith the purpose of the data, which are to be transmitted on atransmission frame. This is to enable the mobile service data to respondswiftly and strongly against noise and channel changes. Thepre-processor 230 may also refer to the transmission parameter whenperforming the additional encoding process. Also, the pre-processor 230groups a plurality of mobile service data packets to configure a datagroup. Thereafter, known data, mobile service data, RS parity data, andMPEG header are allocated to pre-determined regions within the datagroup.

Pre-Processor within Transmitter

FIG. 16 illustrates a block diagram showing the structure of apre-processor 230 according to the present invention. Herein, thepre-processor 230 includes an MPH frame encoder 301, a block processor302, a group formatter 303, a signaling encoder 304, and a packetformatter 305. The MPH frame encoder 301, which is included in thepre-processor 230 having the above-described structure, data-randomizesthe mobile service data that are inputted to the demultiplexer 210,thereby creating a RS frame. Then, the MPH frame encoder 301 performs anencoding process for error correction in RS frame units. The MPH frameencoder 301 may include at least one RS frame encoder. Morespecifically, RS frame encoders may be provided in parallel, wherein thenumber of RS frame encoders is equal to the number of parades within theMPH frame. As described above, the MPH frame is a basic time cycleperiod for transmitting at least one parade. Also, each parade consistsof one or two RS frames.

FIG. 17 illustrates a conceptual block diagram of the MPH frame encoder301 according to an embodiment of the present invention. The MPH frameencoder 301 includes an input demultiplexer (DEMUX) 309, M number of RSframe encoders 310 to 31M-1, and an output multiplexer (MUX) 320.Herein, M represent the number of parades included in one MPH frame. Theinput demultiplexer (DEMUX) 309 splits input ensembles. Then, the splitinput ensembles decide the RS frame to which the ensembles are to beinputted. Thereafter, the inputted ensembles are outputted to therespective RS frame. At this point, an ensemble may be mapped to each RSframe encoder or parade. For example, when one parade configures one RSframe, the ensembles, RS frames, and parades may each be mapped to be ina one-to-one (1:1) correspondence with one another. More specifically,the data in one ensemble configure a RS frame. And, a RS frame isdivided into a plurality of data groups. Based upon the RS frame mode ofTable 1, the data within one RS frame may be assigned either to all ofregions A/B/C/D within multiple data groups, or to at least one ofregions A/B and regions C/D within multiple data groups.

When the RS frame mode value is equal to ‘01’, i.e., when the data ofthe primary RS frame are assigned to regions A/B of the correspondingdata group and data of the secondary RS frame are assigned to regionsC/D of the corresponding data group, each RS frame encoder creates aprimary RS frame and a secondary RS frame for each parade. Conversely,when the RS frame mode value is equal to ‘00’, when the data of theprimary RS frame are assigned to all of regions A/B/C/D, each RS frameencoder creates a RS frame (i.e., a primary RS frame) for each parade.Also, each RS frame encoder divides each RS frame into several portions.Each portion of the RS frame is equivalent to a data amount that can betransmitted by a data group.

The output multiplexer (MUX) 320 multiplexes portions within M number ofRS frame encoders 310 to 310M-1 are multiplexed and then outputted tothe block processor 302. For example, if one parade transmits two RSframes, portions of primary RS frames within M number of RS frameencoders 310 to 310M-1 are multiplexed and outputted. Thereafter,portions of secondary RS frames within M number of RS frame encoders 310to 310M-1 are multiplexed and transmitted. The input demultiplexer(DEMUX) 309 and the output multiplexer (MUX) 320 operate based upon thecontrol of the control unit 200. The control unit 200 may providenecessary (or required) FEC modes to each RS frame encoder. The FEC modeincludes the RS code mode, which will be described in detail in a laterprocess.

FIG. 18 illustrates a detailed block diagram of an RS frame encoderamong a plurality of RS frame encoders within an MPH frame encoder. OneRS frame encoder may include a primary encoder 410 and a secondaryencoder 420. Herein, the secondary encoder 420 may or may not operatebased upon the RS frame mode. For example, when the RS frame mode valueis equal to ‘00’, as shown in Table 1, the secondary encoder 420 doesnot operate. The primary encoder 410 may include a data randomizer 411,a Reed-Solomon-cyclic redundancy check (RS-CRC) encoder (412), and a RSframe divider 413. And, the secondary encoder 420 may also include adata randomizer 421, a RS-CRC encoder (422), and a RS frame divider 423.

More specifically, the data randomizer 411 of the primary encoder 410receives mobile service data of a primary ensemble outputted from theoutput demultiplexer (DEMUX) 309. Then, after randomizing the receivedmobile service data, the data randomizer 411 outputs the randomized datato the RS-CRC encoder 412. At this point, since the data randomizer 411performs the randomizing process on the mobile service data, therandomizing process that is to be performed by the data randomizer 251of the post-processor 250 on the mobile service data may be omitted. Thedata randomizer 411 may also discard the synchronization byte within themobile service data packet and perform the randomizing process. This isan option that may be chosen by the system designer. In the examplegiven in the present invention, the randomizing process is performedwithout discarding the synchronization byte within the correspondingmobile service data packet.

The RS-CRC encoder 412 uses at least one of a Reed-Solomon (RS) code anda cyclic redundancy check (CRC) code, so as to perform forward errorcollection (FEC) encoding on the randomized primary ensemble, therebyforming a primary RS frame. Therefore, the RS-CRC encoder 412 outputsthe newly formed primary RS frame to the RS frame divider 413. TheRS-CRC encoder 412 groups a plurality of mobile service data packetsthat is randomized and inputted, so as to create a RS frame. Then, theRS-CRC encoder 412 performs at least one of an error correction encodingprocess and an error detection encoding process in RS frame units.Accordingly, robustness may be provided to the mobile service data,thereby scattering group error that may occur during changes in afrequency environment, thereby enabling the mobile service data torespond to the frequency environment, which is extremely vulnerable andliable to frequent changes. Also, the RS-CRC encoder 412 groups aplurality of RS frame so as to create a super frame, thereby performinga row permutation process in super frame units. The row permutationprocess may also be referred to as a “row interleaving process”.Hereinafter, the process will be referred to as “row permutation” forsimplicity.

More specifically, when the RS-CRC encoder 412 performs the process ofpermuting each row of the super frame in accordance with apre-determined rule, the position of the rows within the super framebefore and after the row permutation process is changed. If the rowpermutation process is performed by super frame units, and even thoughthe section having a plurality of errors occurring therein becomes verylong, and even though the number of errors included in the RS frame,which is to be decoded, exceeds the extent of being able to becorrected, the errors become dispersed within the entire super frame.Thus, the decoding ability is even more enhanced as compared to a singleRS frame.

At this point, as an example of the present invention, RS-encoding isapplied for the error correction encoding process, and a cyclicredundancy check (CRC) encoding is applied for the error detectionprocess in the RS-CRC encoder 412. When performing the RS-encoding,parity data that are used for the error correction are generated. And,when performing the CRC encoding, CRC data that are used for the errordetection are generated. The CRC data generated by CRC encoding may beused for indicating whether or not the mobile service data have beendamaged by the errors while being transmitted through the channel. Inthe present invention, a variety of error detection coding methods otherthan the CRC encoding method may be used, or the error correction codingmethod may be used to enhance the overall error correction ability ofthe receiving system. Herein, the RS-CRC encoder 412 refers to apre-determined transmission parameter provided by the control unit 200and/or a transmission parameter provided from the service multiplexer100 so as to perform operations including RS frame configuration, RSencoding, CRC encoding, super frame configuration, and row permutationin super frame units.

FIG. 19 illustrates a process of one or two RS frame being divided intoseveral portions, based upon an RS frame mode value, and a process ofeach portion being assigned to a corresponding region within therespective data group. More specifically, FIG. 19( a) shows an exampleof the RS frame mode value being equal to ‘00’. Herein, only the primaryencoder 410 of FIG. 18 operates, thereby forming one RS frame for oneparade. Then, the RS frame is divided into several portions, and thedata of each portion are assigned to regions A/B/C/D within therespective data group. FIG. 19( b) shows an example of the RS frame modevalue being equal to ‘01’. Herein, both the primary encoder 410 and thesecondary encoder 420 of FIG. 18 operate, thereby forming two RS framesfor one parade, i.e., one primary RS frame and one secondary RS frame.Then, the primary RS frame is divided into several portions, and thesecondary RS frame is divided into several portions. At this point, thedata of each portion of the primary RS frame are assigned to regions A/Bwithin the respective data group. And, the data of each portion of thesecondary RS frame are assigned to regions C/D within the respectivedata group.

Detailed Description of the RS Frame

FIG. 20( a) illustrates an example of an RS frame being generated fromthe RS-CRC encoder 412 according to the present invention. According tothis embodiment, in the RS frame, the length of a column (i.e., numberof rows) is set to 187 bytes, and the length of a row (i.e., number ofcolumn) is set to N bytes. At this point, the value of N, whichcorresponds to the number of columns within an RS frame, can be decidedaccording to Equation 2.

$\begin{matrix}{N = {\left\lfloor \frac{5 \times {NoG} \times {PL}}{187 + P} \right\rfloor - 2}} & {{Equation}\mspace{14mu} 2}\end{matrix}$

Herein, NoG indicates the number of data groups assigned to a sub-frame.PL represents the number of SCCC payload data bytes assigned to a datagroup. And, P signifies the number of RS parity data bytes added to eachcolumn of the RS frame. Finally, └X┘ is the greatest integer that isequal to or smaller than X.

More specifically, in Equation 2, PL corresponds to the length of an RSframe portion. The value of PL is equivalent to the number of SCCCpayload data bytes that are assigned to the corresponding data group.Herein, the value of PL may vary depending upon the RS frame mode, SCCCblock mode, and SCCC outer code mode. Table 2 to Table 5 belowrespectively show examples of PL values, which vary in accordance withthe RS frame mode, SCCC block mode, and SCCC outer code mode. The SCCCblock mode and the SCCC outer code mode will be described in detail in alater process.

TABLE 2 SCCC outer code mode for Region A for Region B for Region C forRegion D PL 00 00 00 00 9624 00 00 00 01 9372 00 00 01 00 8886 00 00 0101 8634 00 01 00 00 8403 00 01 00 01 8151 00 01 01 00 7665 00 01 01 017413 01 00 00 00 7023 01 00 00 01 6771 01 00 01 00 6285 01 00 01 01 603301 01 00 00 5802 01 01 00 01 5550 01 01 01 00 5064 01 01 01 01 4812Others Reserved

Table 2 shows an example of the PL values for each data group within anRS frame, wherein each PL value varies depending upon the SCCC outercode mode, when the RS frame mode value is equal to ‘00’, and when theSCCC block mode value is equal to ‘00’. For example, when it is assumedthat each SCCC outer code mode value of regions A/B/C/D within the datagroup is equal to ‘00’ (i.e., the block processor 302 of a later blockperforms encoding at a coding rate of ½), the PL value within each datagroup of the corresponding RS frame may be equal to 9624 bytes. Morespecifically, 9624 bytes of mobile service data within one RS frame maybe assigned to regions A/B/C/D of the corresponding data group.

TABLE 3 SCCC outer code mode PL 00 9624 01 4812 Others Reserved

Table 3 shows an example of the PL values for each data group within anRS frame, wherein each PL value varies depending upon the SCCC outercode mode, when the RS frame mode value is equal to ‘00’, and when theSCCC block mode value is equal to ‘01’.

TABLE 4 SCCC outer code mode for Region A for Region B PL 00 00 7644 0001 6423 01 00 5043 01 01 3822 Others Reserved

Table 4 shows an example of the PL values for each data group within aprimary RS frame, wherein each PL value varies depending upon the SCCCouter code mode, when the RS frame mode value is equal to ‘01’, and whenthe SCCC block mode value is equal to ‘00’. For example, when each SCCCouter code mode value of regions A/B is equal to ‘00’, 7644 bytes ofmobile service data within a primary RS frame may be assigned to regionsA/B of the corresponding data group.

TABLE 5 SCCC outer code mode for Region C for Region D PL 00 00 1980 0001 1728 01 00 1242 01 01 990 Others Reserved

Table 5 shows an example of the PL values for each data group within asecondary RS frame, wherein each PL value varies depending upon the SCCCouter code mode, when the RS frame mode value is equal to ‘01’, and whenthe SCCC block mode value is equal to ‘00’. For example, when each SCCCouter code mode value of regions C/D is equal to ‘00’, 1980 bytes ofmobile service data within a secondary RS frame may be assigned toregions C/D of the corresponding data group.

According to the embodiment of the present invention, the value of N isequal to or greater than 187 (i.e., N≧187). More specifically, the RSframe of FIG. 20( a) has the size of N(row)×187(column) bytes. Morespecifically, the RS-CRC encoder 412 first divides the inputted mobileservice data bytes to units of a predetermined length. The predeterminedlength is decided by the system designer. And, in the example of thepresent invention, the predetermined length is equal to 187 bytes, and,therefore, the 187-byte unit will be referred to as a “packet” forsimplicity. For example, the inputted mobile service data may correspondeither to an MPEG transport stream (TS) packet configured of 188-byteunits or to an IP datagram. Alternatively, the IP datagram may beencapsulated to a TS packet of 188-byte units and, then, inputted.

When the mobile service data that are being inputted correspond to aMPEG transport packet stream configured of 188-byte units, the firstsynchronization byte is removed so as to configure a 187-byte unit.Then, N number of packets are grouped to form an RS frame. Herein, thesynchronization byte is removed because each mobile service data packethas the same value. Meanwhile, when the input mobile service data of theRS frame do not correspond to the MPEG TS packet format, the mobileservice data are inputted N number of times in 187-byte units withoutbeing processed with the removing of the MPEG synchronization byte,thereby creating a RS frame.

In addition, when the input data format of the RS frame supports boththe input data corresponding to the MPEG TS packet and the input datanot corresponding to the MPEG TS packet, such information may beincluded in a transmission parameter transmitted from the servicemultiplexer 100, thereby being sent to the transmitter 200. Accordingly,the RS-CRC encoder 412 of the transmitter 200 receives this informationto be able to control whether or not to perform the process of removingthe MPEG synchronization byte. Also, the transmitter provides suchinformation to the receiving system so as to control the process ofinserting the MPEG synchronization byte that is to be performed by theRS frame decoder of the receiving system. Herein, the process ofremoving the synchronization byte may be performed during a randomizingprocess of the data randomizer 411 in an earlier process. In this case,the process of the removing the synchronization byte by the RS-CRCencoder 412 may be omitted.

Moreover, when adding synchronization bytes from the receiving system,the process may be performed by the data derandomizer instead of the RSframe decoder. Therefore, if a removable fixed byte (e.g.,synchronization byte) does not exist within the mobile service datapacket that is being inputted to the RS-CRC encoder 412, or if themobile service data that are being inputted are not configured in apacket format, the mobile service data that are being inputted aredivided into 187-byte units, thereby configuring a packet for each187-byte unit.

Subsequently, N number of packets configured of 187 bytes is grouped toconfigure a RS frame. At this point, the RS frame is configured as a RSframe having the size of N(row)×187(column) bytes, in which 187-bytepackets are sequentially inputted in a row direction. More specifically,each of the N number of columns included in the RS frame includes 187bytes. When the RS frame is created, as shown in FIG. 20( a), the RS-CRCencoder 412 performs a (Nc,Kc)−RS encoding process on each column, so asto generate Nc−Kc(=P) number of parity bytes. Then, the RS-CRC encoder412 adds the newly generated P number of parity bytes after the verylast byte of the corresponding column, thereby creating a column of(187+P) bytes. Herein, as shown in FIG. 20( a), Kc is equal to 187(i.e., Kc=187), and Nc is equal to 187+P (i.e., Nc=187+P). Herein, thevalue of P may vary depending upon the RS code mode. Table 6 below showsan example of an RS code mode, as one of the RS encoding information.

TABLE 6 RS code mode RS code Number of Parity Bytes (P) 00 (211, 187) 2401 (223, 187) 36 10 (235, 187) 48 11 Reserved Reserved

Table 6 shows an example of 2 bits being assigned in order to indicatethe RS code mode. The RS code mode represents the number of parity bytescorresponding to the RS frame. For example, when the RS code mode valueis equal to ‘10’, (235,187)-RS-encoding is performed on the RS frame ofFIG. 20( a), so as to generate 48 parity data bytes. Thereafter, the 48parity bytes are added after the last data byte of the correspondingcolumn, thereby creating a column of 235 data bytes. When the RS framemode value is equal to ‘00’ in Table 1 (i.e., when the RS frame modeindicates a single RS frame), only the RS code mode of the correspondingRS frame is indicated. However, when the RS frame mode value is equal to‘01’ in Table 1 (i.e., when the RS frame mode indicates multiple RSframes), the RS code mode corresponding to a primary RS frame and asecondary RS frame. More specifically, it is preferable that the RS codemode is independently applied to the primary RS frame and the secondaryRS frame.

When such RS encoding process is performed on all N number of columns, aRS frame having the size of N(row)×(187+P)(column) bytes may be created,as shown in FIG. 20( b). Each row of the RS frame is configured of Nbytes. However, depending upon channel conditions between thetransmitting system and the receiving system, error may be included inthe RS frame. When errors occur as described above, CRC data (or CRCcode or CRC checksum) may be used on each row unit in order to verifywhether error exists in each row unit. The RS-CRC encoder 412 mayperform CRC encoding on the mobile service data being RS encoded so asto create (or generate) the CRC data. The CRC data being generated byCRC encoding may be used to indicate whether the mobile service datahave been damaged while being transmitted through the channel.

The present invention may also use different error detection encodingmethods other than the CRC encoding method. Alternatively, the presentinvention may use the error correction encoding method to enhance theoverall error correction ability of the receiving system. FIG. 20( c)illustrates an example of using a 2-byte (i.e., 16-bit) CRC checksum asthe CRC data. Herein, a 2-byte CRC checksum is generated for N number ofbytes of each row, thereby adding the 2-byte CRC checksum at the end ofthe N number of bytes. Thus, each row is expanded to (N+2) number ofbytes. Equation 3 below corresponds to an exemplary equation forgenerating a 2-byte CRC checksum for each row being configured of Nnumber of bytes.

g(x)=x ¹⁶ +x ¹² +x ⁵+1  Equation 3

The process of adding a 2-byte checksum in each row is only exemplary.Therefore, the present invention is not limited only to the exampleproposed in the description set forth herein. As described above, whenthe process of RS encoding and CRC encoding are completed, the(N×187)-byte RS frame is expanded to a (N+2)×(187+P)-byte RS frame.Based upon an error correction scenario of a RS frame expanded asdescribed above, the data bytes within the RS frame are transmittedthrough a channel in a row direction. At this point, when a large numberof errors occur during a limited period of transmission time, errorsalso occur in a row direction within the RS frame being processed with adecoding process in the receiving system. However, in the perspective ofRS encoding performed in a column direction, the errors are shown asbeing scattered. Therefore, error correction may be performed moreeffectively. At this point, a method of increasing the number of paritydata bytes (P) may be used in order to perform a more intense errorcorrection process. However, using this method may lead to a decrease intransmission efficiency. Therefore, a mutually advantageous method isrequired. Furthermore, when performing the decoding process, an erasuredecoding process may be used to enhance the error correctionperformance.

Additionally, the RS-CRC encoder 412 according to the present inventionalso performs a row permutation (or interleaving) process in super frameunits in order to further enhance the error correction performance whenerror correction the RS frame. FIG. 21( a) to FIG. 21( d) illustrates anexample of performing a row permutation process in super frame unitsaccording to the present invention. More specifically, G number of RSframes RS-CRC-encoded is grouped to form a super frame, as shown in FIG.21( a). At this point, since each RS frame is formed of (N+2)×(187+P)number of bytes, one super frame is configured to have the size of(N+2)×(187+P)×G bytes.

When a row permutation process permuting each row of the super frameconfigured as described above is performed based upon a pre-determinedpermutation rule, the positions of the rows prior to and after beingpermuted (or interleaved) within the super frame may be altered. Morespecifically, the i^(th) row of the super frame prior to theinterleaving process, as shown in FIG. 21( b), is positioned in thej^(th) row of the same super frame after the row permutation process, asshown in FIG. 21( c). The above-described relation between i and j canbe easily understood with reference to a permutation rule as shown inEquation 4 below.

j=G(imod(187+P))+└i/(187+P)┘

i=(187+P)(jmodG)+└j/G┘  Equation 4

where 0≦i, j≦(187+P)G−1; or

where 0≦i, j<(187+P)G

Herein, each row of the super frame is configured of (N+2) number ofdata bytes even after being row-permuted in super frame units.

When all row permutation processes in super frame units are completed,the super frame is once again divided into G number of row-permuted RSframes, as shown in FIG. 21( d), and then provided to the RS framedivider 413. Herein, the number of RS parity bytes and the number ofcolumns should be equally provided in each of the RS frames, whichconfigure a super frame. As described in the error correction scenarioof a RS frame, in case of the super frame, a section having a largenumber of error occurring therein is so long that, even when one RSframe that is to be decoded includes an excessive number of errors(i.e., to an extent that the errors cannot be corrected), such errorsare scattered throughout the entire super frame. Therefore, incomparison with a single RS frame, the decoding performance of the superframe is more enhanced.

The above description of the present invention corresponds to theprocesses of forming (or creating) and encoding an RS frame, when a datagroup is divided into regions A/B/C/D, and when data of an RS frame areassigned to all of regions A/B/C/D within the corresponding data group.More specifically, the above description corresponds to an embodiment ofthe present invention, wherein one RS frame is transmitted using oneparade. In this embodiment, the secondary encoder 420 does not operate(or is not active).

Meanwhile, 2 RS frames are transmitting using one parade, the data ofthe primary RS frame may be assigned to regions A/B within the datagroup and be transmitted, and the data of the secondary RS frame may beassigned to regions C/D within the data group and be transmitted. Atthis point, the primary encoder 410 receives the mobile service datathat are to be assigned to regions A/B within the data group, so as toform the primary RS frame, thereby performing RS-encoding andCRC-encoding. Similarly, the secondary encoder 420 receives the mobileservice data that are to be assigned to regions C/D within the datagroup, so as to form the secondary RS frame, thereby performingRS-encoding and CRC-encoding. More specifically, the primary RS frameand the secondary RS frame are created independently.

FIG. 22 illustrates examples of receiving the mobile service data thatare to be assigned to regions A/B within the data group, so as to formthe primary RS frame, and receives the mobile service data that are tobe assigned to regions C/D within the data group, so as to form thesecondary RS frame, thereby performing error correction encoding anderror detection encoding on each of the first and secondary RS frames.More specifically, FIG. 22( a) illustrates an example of the RS-CRCencoder 412 of the primary encoder 410 receiving mobile service data ofthe primary ensemble that are to be assigned to regions A/B within thecorresponding data group, so as to create an RS frame having the size ofN1(row)×187(column). Then, in this example, the primary encoder 410performs RS-encoding on each column of the RS frame created as describedabove, thereby adding P1 number of parity data bytes in each column.Finally, the primary encoder 410 performs CRC-encoding on each row,thereby adding a 2-byte checksum in each row.

FIG. 22( b) illustrates an example of the RS-CRC encoder 422 of thesecondary encoder 420 receiving mobile service data of the secondaryensemble that are to be assigned to regions C/D within the correspondingdata group, so as to create an RS frame having the size ofN2(row)×187(column). Then, in this example, the secondary encoder 420performs RS-encoding on each column of the RS frame created as describedabove, thereby adding P2 number of parity data bytes in each column.Finally, the secondary encoder 420 performs CRC-encoding on each row,thereby adding a 2-byte checksum in each row. At this point, each of theRS-CRC encoders 412 and 422 may refer to a pre-determined transmissionparameter provided by the control unit 200 and/or a transmissionparameter provided from the service multiplexer 100, the RS-CRC encoders412 and 422 may be informed of RS frame information (including RS framemode), RS encoding information (including RS code mode), SCCCinformation (including SCCC block information and SCCC outer code mode),data group information, and region information within a data group. TheRS-CRC encoders 412 and 422 may refer to the transmission parameters forthe purpose of RS frame configuration, error correction encoding, errordetection encoding. Furthermore, the transmission parameters should alsobe transmitted to the receiving system so that the receiving system canperform a normal decoding process.

The data of the primary RS frame, which is encoded by RS frame units androw-permuted by super frame units from the RS-CRC encoder 412 of theprimary encoder 410, are outputted to the RS frame divider 413. If thesecondary encoder 420 also operates in the embodiment of the presentinvention, the data of the secondary RS frame, which is encoded by RSframe units and row-permuted by super frame units from the RS-CRCencoder 422 of the secondary encoder 420, are outputted to the RS framedivider 423. The RS frame divider 413 of the primary encoder 410 dividesthe primary RS frame into several portions, which are then outputted tothe output multiplexer (MUX) 320. Each portion of the primary RS frameis equivalent to a data amount that can be transmitted by one datagroup. Similarly, the RS frame divider 423 of the secondary encoder 420divides the secondary RS frame into several portions, which are thenoutputted to the output multiplexer (MUX) 320.

Hereinafter, the RS frame divider 413 of the primary RS encoder 410 willnow be described in detail. Also, in order to simplify the descriptionof the present invention, it is assumed that an RS frame having the sizeof N(row)×187(column), as shown in FIG. 20( a) to FIG. 20( c), that Pnumber of parity data bytes are added to each column by RS-encoding theRS frame, and that a 2-byte checksum is added to each row byCRC-encoding the RS frame. Accordingly, the RS frame divider 413 divides(or partitions) the encoded RS frame having the size of(N+2)(row)×187(column) into several portions, each having the size of PL(wherein PL corresponds to the length of the RS frame portion).

At this point, as shown in Table 2 to Table 5, the value of PL may varydepending upon the RS frame mode, SCCC block mode, and SCCC outer codermode. Also, the total number of data bytes of the RS-encoded andCRC-encoded RS frame is equal to or smaller than 5×NoG×PL. In this case,the RS frame is divided (or partitioned) into ((5×NoG)−1) number ofportions each having the size of PL and one portion having a size equalto smaller than PL. More specifically, with the exception of the lastportion of the RS frame, each of the remaining portions of the RS framehas an equal size of PL. If the size of the last portion is smaller thanPL, a stuffing byte (or dummy byte) may be inserted in order to fill (orreplace) the lacking number of data bytes, thereby enabling the lastportion of the RS frame to also be equal to PL. Each portion of an RSframe corresponds to the amount of data that are to be SCCC-encoded andmapped into a single data group of a parade.

FIG. 23( a) and FIG. 23( b) respectively illustrate examples of adding Snumber of stuffing bytes, when an RS frame having the size of(N+2)(row)×(187+P)(column) is divided into 5×NoG number of portions,each having the size of PL. More specifically, the RS-encoded andCRC-encoded RS frame, shown in FIG. 23( a), is divided into severalportions, as shown in FIG. 23( b). The number of divided portions at theRS frame is equal to (5×NoG). Particularly, the first ((5×NoG)−1) numberof portions each has the size of PL, and the last portion of the RSframe may be equal to or smaller than PL. If the size of the lastportion is smaller than PL, a stuffing byte (or dummy byte) may beinserted in order to fill (or replace) the lacking number of data bytes,as shown in Equation 5 below, thereby enabling the last portion of theRS frame to also be equal to PL.

S=(5×NoG×PL)−((N+2)×(187+P))  Equation 5

Herein, each portion including data having the size of PL passes throughthe output multiplexer 320 of the MPH frame encoder 301, which is thenoutputted to the block processor 302.

At this point, the mapping order of the RS frame portions to a parade ofdata groups in not identical with the group assignment order defined inEquation 1. When given the group positions of a parade in an MPH frame,the SCCC-encoded RS frame portions will be mapped in a time order (i.e.,in a left-to-right direction). For example, as shown in FIG. 11, datagroups of the 2^(nd) parade (Parade #1) are first assigned (orallocated) to the 13^(th) slot (Slot #12) and then assigned to the3^(rd) slot (Slot #2). However, when the data are actually placed in theassigned slots, the data are placed in a time sequence (or time order,i.e., in a left-to-right direction). More specifically, the 1^(st) datagroup of Parade #1 is placed in Slot #2, and the 2^(nd) data group ofParade #1 is placed in Slot #12.

Block Processor

Meanwhile, the block processor 302 performs an SCCC outer encodingprocess on the output of the MPH frame encoder 301. More specifically,the block processor 302 receives the data of each error correctionencoded portion. Then, the block processor 302 encodes the data onceagain at a coding rate of 1/H (wherein H is an integer equal to orgreater than 2 (i.e., H≧22)), thereby outputting the 1/H-rate encodeddata to the group formatter 303. According to the embodiment of thepresent invention, the input data are encoded either at a coding rate of½ (also referred to as “½-rate encoding”) or at a coding rate of ¼ (alsoreferred to as “¼-rate encoding”). The data of each portion outputtedfrom the MPH frame encoder 301 may include at least one of pure mobileservice data, RS parity data, CRC data, and stuffing data. However, in abroader meaning, the data included in each portion may correspond todata for mobile services. Therefore, the data included in each portionwill all be considered as mobile service data and described accordingly.

The group formatter 303 inserts the mobile service dataSCCC-outer-encoded and outputted from the block processor 302 in thecorresponding region within the data group, which is formed inaccordance with a pre-defined rule. Also, in association with the datadeinterleaving process, the group formatter 303 inserts various placeholders (or known data place holders) in the corresponding region withinthe data group. Thereafter, the group formatter 303 deinterleaves thedata within the data group and the place holders.

According to the present invention, with reference to data after beingdata-interleaved, as shown in FIG. 5, a data groups is configured of 10MPH blocks (B1 to B10) and divided into 4 regions (A, B, C, and D).Also, as shown in FIG. 5, when it is assumed that the data group isdivided into a plurality of hierarchical regions, as described above,the block processor 302 may encode the mobile service data, which are tobe inserted to each region based upon the characteristic of eachhierarchical region, at different coding rates. For example, the blockprocessor 302 may encode the mobile service data, which are to beinserted in region A/B within the corresponding data group, at a codingrate of ½. Then, the group formatter 303 may insert the ½-rate encodedmobile service data to region A/B. Also, the block processor 302 mayencode the mobile service data, which are to be inserted in region C/Dwithin the corresponding data group, at a coding rate of ¼ having higher(or stronger) error correction ability than the ½-coding rate.Thereafter, the group formatter 303 may insert the ½-rate encoded mobileservice data to region C/D. In another example, the block processor 302may encode the mobile service data, which are to be inserted in regionC/D, at a coding rate having higher error correction ability than the¼-coding rate. Then, the group formatter 303 may either insert theencoded mobile service data to region C/D, as described above, or leavethe data in a reserved region for future usage.

According to another embodiment of the present invention, the blockprocessor 302 may perform a 1/H-rate encoding process in SCCC blockunits. Herein, the SCCC block includes at least one MPH block. At thispoint, when 1/H-rate encoding is performed in MPH block units, the MPHblocks (B1 to B10) and the SCCC block (SCB1 to SCB10) become identicalto one another (i.e., SCB1=B1, SCB2=B2, SCB3=B3, SCB4=B4, SCB5=B5,SCB6=B6, SCB7=B7, SCB8=B8, SCB9=B9, and SCB10=B10). For example, the MPHblock 1 (B1) may be encoded at the coding rate of ½, the MPH block 2(B2) may be encoded at the coding rate of ¼, and the MPH block 3 (B3)may be encoded at the coding rate of ½. The coding rates are appliedrespectively to the remaining MPH blocks.

Alternatively, a plurality of MPH blocks within regions A, B, C, and Dmay be grouped into one SCCC block, thereby being encoded at a codingrate of 1/H in SCCC block units. Accordingly, the receiving performanceof region C/D may be enhanced. For example, MPH block 1 (B1) to MPHblock 5 (B5) may be grouped into one SCCC block and then encoded at acoding rate of ½. Thereafter, the group formatter 303 may insert the½-rate encoded mobile service data to a section starting from MPH block1 (B1) to MPH block 5 (B5). Furthermore, MPH block 6 (B6) to MPH block10 (B10) may be grouped into one SCCC block and then encoded at a codingrate of ¼. Thereafter, the group formatter 303 may insert the ¼-rateencoded mobile service data to another section starting from MPH block 6(B6) to MPH block 10 (B10). In this case, one data group may consist oftwo SCCC blocks.

According to another embodiment of the present invention, one SCCC blockmay be formed by grouping two MPH blocks. For example, MPH block 1 (B1)and MPH block 6 (B6) may be grouped into one SCCC block (SCB1).Similarly, MPH block 2 (B2) and MPH block 7 (B7) may be grouped intoanother SCCC block (SCB2). Also, MPH block 3 (B3) and MPH block 8 (B8)may be grouped into another SCCC block (SCB3). And, MPH block 4 (B4) andMPH block 9 (B9) may be grouped into another SCCC block (SCB4).Furthermore, MPH block 5 (B5) and MPH block 10 (B10) may be grouped intoanother SCCC block (SCB5). In the above-described example, the datagroup may consist of 10 MPH blocks and 5 SCCC blocks. Accordingly, in adata (or signal) receiving environment undergoing frequent and severechannel changes, the receiving performance of regions C and D, which isrelatively more deteriorated than the receiving performance of region A,may be reinforced. Furthermore, since the number of mobile service datasymbols increases more and more from region A to region D, the errorcorrection encoding performance becomes more and more deteriorated.Therefore, when grouping a plurality of MPH block to form one SCCCblock, such deterioration in the error correction encoding performancemay be reduced.

As described-above, when the block processor 302 performs encoding at a1/H-coding rate, information associated with SCCC should be transmittedto the receiving system in order to accurately recover the mobileservice data. Table 7 below shows an example of a SCCC block mode, whichindicating the relation between an MPH block and an SCCC block, amongdiverse SCCC block information.

TABLE 7 SCCC Block Mode 00 01 10 11 Description One MPH Block Two MPHBlocks per SCCC Block per SCCC Block SCB input, SCB input, SCB MPH BlockMPH Blocks Reserved Reserved SCB1 B1 B1 + B6 SCB2 B2 B2 + B7 SCB3 B3B3 + B8 SCB4 B4 B4 + B9 SCB5 B5 B5 + B10 SCB6 B6 — SCB7 B7 — SCB8 B8 —SCB9 B9 — SCB10 B10 —

More specifically, Table 4 shows an example of 2 bits being allocated inorder to indicate the SCCC block mode. For example, when the SCCC blockmode value is equal to ‘00’, this indicates that the SCCC block and theMPH block are identical to one another. Also, when the SCCC block modevalue is equal to ‘01’, this indicates that each SCCC block isconfigured of 2 MPH blocks.

As described above, if one data group is configured of 2 SCCC blocks,although it is not indicated in Table 7, this information may also beindicated as the SCCC block mode. For example, when the SCCC block modevalue is equal to ‘10’, this indicates that each SCCC block isconfigured of 5 MPH blocks and that one data group is configured of 2SCCC blocks. Herein, the number of MPH blocks included in an SCCC blockand the position of each MPH block may vary depending upon the settingsmade by the system designer. Therefore, the present invention will notbe limited to the examples given herein. Accordingly, the SCCC modeinformation may also be expanded.

An example of a coding rate information of the SCCC block, i.e., SCCCouter code mode, is shown in Table 8 below.

TABLE 8 SCCC outer code mode (2 bits) Description 00 Outer code rate ofSCCC block is ½ rate 01 Outer code rate of SCCC block is ¼ rate 10Reserved 11 Reserved

More specifically, Table 8 shows an example of 2 bits being allocated inorder to indicate the coding rate information of the SCCC block. Forexample, when the SCCC outer code mode value is equal to ‘00’, thisindicates that the coding rate of the corresponding SCCC block is ½.And, when the SCCC outer code mode value is equal to ‘01’, thisindicates that the coding rate of the corresponding SCCC block is ¼.

If the SCCC block mode value of Table 7 indicates ‘00’, the SCCC outercode mode may indicate the coding rate of each MPH block with respect toeach MPH block. In this case, since it is assumed that one data groupincludes 10 MPH blocks and that 2 bits are allocated for each SCCC blockmode, a total of 20 bits are required for indicating the SCCC blockmodes of the 10 MPH modes. In another example, when the SCCC block modevalue of Table 7 indicates ‘00’, the SCCC outer code mode may indicatethe coding rate of each region with respect to each region within thedata group. In this case, since it is assumed that one data groupincludes 4 regions (i.e., regions A, B, C, and D) and that 2 bits areallocated for each SCCC block mode, a total of 8 bits are required forindicating the SCCC block modes of the 4 regions. In another example,when the SCCC block mode value of Table 7 is equal to ‘01’, each of theregions A, B, C, and D within the data group has the same SCCC outercode mode.

Meanwhile, an example of an SCCC output block length (SOBL) for eachSCCC block, when the SCCC block mode value is equal to ‘00’, is shown inTable 9 below.

TABLE 9 SIBL SCCC Block SOBL ½ rate ¼ rate SCB1 (B1) 528 264 132 SCB2(B2) 1536 768 384 SCB3 (B3) 2376 1188 594 SCB4 (B4) 2388 1194 597 SCB5(B5) 2772 1386 693 SCB6 (B6) 2472 1236 618 SCB7 (B7) 2772 1386 693 SCB8(B8) 2508 1254 627 SCB9 (B9) 1416 708 354 SCB10 (B10) 480 240 120

More specifically, when given the SCCC output block length (SOBL) foreach SCCC block, an SCCC input block length (SIBL) for eachcorresponding SCCC block may be decided based upon the outer coding rateof each SCCC block. The SOBL is equivalent to the number of SCCC output(or outer-encoded) bytes for each SCCC block. And, the SIBL isequivalent to the number of SCCC input (or payload) bytes for each SCCCblock. Table 10 below shows an example of the SOBL and SIBL for eachSCCC block, when the SCCC block mode value is equal to ‘01’.

TABLE 10 SIBL SCCC Block SOBL ½ rate ¼ rate SCB1 (B1 + B6) 528 264 132SCB2 (B2 + B7) 1536 768 384 SCB3 (B3 + B8) 2376 1188 594 SCB4 (B4 + B9)2388 1194 597 SCB5 (B5 + B10) 2772 1386 693

In order to do so, as shown in FIG. 24, the block processor 302 includesa RS frame portion-SCCC block converter 511, a byte-bit converter 512, aconvolution encoder 513, a symbol interleaver 514, a symbol-byteconverter 515, and an SCCC block-MPH block converter 516. Theconvolutional encoder 513 and the symbol interleaver 514 are virtuallyconcatenated with the trellis encoding module in the post-processor inorder to configure an SCCC block. More specifically, the RS frameportion-SCCC block converter 511 divides the RS frame portions, whichare being inputted, into multiple SCCC blocks using the SIBL of Table 9and Table 10 based upon the RS code mode, SCCC block mode, and SCCCouter code mode. Herein, the MPH frame encoder 301 may output onlyprimary RS frame portions or both primary RS frame portions andsecondary RS frame portions in accordance with the RS frame mode.

When the RS Frame mode is set to ‘00’, a portion of the primary RS Frameequal to the amount of data, which are to be SCCC outer encoded andmapped to 10 MPH blocks (B1 to B10) of a data group, will be provided tothe block processor 302. When the SCCC block mode value is equal to‘00’, then the primary RS frame portion will be split into 10 SCCCBlocks according to Table 9. Alternatively, when the SCCC block modevalue is equal to ‘01’, then the primary RS frame will be split into 5SCCC blocks according to Table 10.

When the RS frame mode value is equal to ‘01’, then the block processor302 may receive two RS frame portions. The RS frame mode value of ‘01’will not be used with the SCCC block mode value of ‘01’. The firstportion from the primary RS frame will be SCCC-outer-encoded as SCCCBlocks SCB3, SCB4, SCB5, SCB6, SCB7, and SCB8 by the block processor302. The SCCC Blocks S3 and S8 will be mapped to region B and the SCCCblocks SCB4, SCB5, SCB6, and SCB7 shall be mapped to region A by thegroup formatter 303. The second portion from the secondary RS frame willalso be SCCC-outer-encoded, as SCB1, SCB2, SCB9, and SCB10, by the blockprocessor 302. The group formatter 303 will map the SCCC blocks SCB1 andSCB10 to region D as the MPH blocks B1 and B10, respectively. Similarly,the SCCC blocks SCB2 and SCB9 will be mapped to region C as the MPHblocks B2 and B9.

The byte-bit converter 512 identifies the mobile service data bytes ofeach SCCC block outputted from the RS frame portion-SCCC block converter511 as data bits, which are then outputted to the convolution encoder513. The convolution encoder 513 performs one of ½-rate encoding and¼-rate encoding on the inputted mobile service data bits.

FIG. 25 illustrates a detailed block diagram of the convolution encoder513. The convolution encoder 513 includes two delay units 521 and 523and three adders 522, 524, and 525. Herein, the convolution encoder 513encodes an input data bit U and outputs the coded bit U to 5 bits (u0 tou4). At this point, the input data bit U is directly outputted asuppermost bit u0 and simultaneously encoded as lower bit u1u2u3u4 andthen outputted. More specifically, the input data bit U is directlyoutputted as the uppermost bit u0 and simultaneously outputted to thefirst and third adders 522 and 525.

The first adder 522 adds the input data bit U and the output bit of thefirst delay unit 521 and, then, outputs the added bit to the seconddelay unit 523. Then, the data bit delayed by a pre-determined time(e.g., by 1 clock) in the second delay unit 523 is outputted as a lowerbit u1 and simultaneously fed-back to the first delay unit 521. Thefirst delay unit 521 delays the data bit fed-back from the second delayunit 523 by a pre-determined time (e.g., by 1 clock). Then, the firstdelay unit 521 outputs the delayed data bit as a lower bit u2 and, atthe same time, outputs the fed-back data to the first adder 522 and thesecond adder 524. The second adder 524 adds the data bits outputted fromthe first and second delay units 521 and 523 and outputs the added databits as a lower bit u3. The third adder 525 adds the input data bit Uand the output of the second delay unit 523 and outputs the added databit as a lower bit u4.

At this point, the first and second delay units 521 and 523 are reset to‘0’, at the starting point of each SCCC block. The convolution encoder513 of FIG. 25 may be used as a ½-rate encoder or a ¼-rate encoder. Morespecifically, when a portion of the output bit of the convolutionencoder 513, shown in FIG. 25, is selected and outputted, theconvolution encoder 513 may be used as one of a ½-rate encoder and a¼-rate encoder. Table 11 below shown an example of output symbols of theconvolution encoder 513.

TABLE 11 ¼ rate SCCC block SCCC block Region ½ rate mode = ‘00’ mode =‘01’ A, B (u0, u1) (u0, u2), (u1, u3) (u0, u2), (u1, u4) C, D (u0, u1),(u3, u4)

For example, at the ½-coding rate, 1 output symbol (i.e., u0 and u1bits) may be selected and outputted. And, at the ¼-coding rate,depending upon the SCCC block mode, 2 output symbols (i.e., 4 bits) maybe selected and outputted. For example, when the SCCC block mode valueis equal to ‘01’, and when an output symbol configured of u0 and u2 andanother output symbol configured of u1 and u4 are selected andoutputted, a ¼-rate coding result may be obtained.

The mobile service data encoded at the coding rate of ½ or ¼ by theconvolution encoder 513 are outputted to the symbol interleaver 514. Thesymbol interleaver 514 performs block interleaving, in symbol units, onthe output data symbol of the convolution encoder 513. Morespecifically, the symbol interleaver 514 is a type of block interleaver.Any interleaver performing structural rearrangement (or realignment) maybe applied as the symbol interleaver 514 of the block processor.However, in the present invention, a variable length symbol interleaverthat can be applied even when a plurality of lengths is provided for thesymbol, so that its order may be rearranged, may also be used.

FIG. 26 illustrates a symbol interleaver according to an embodiment ofthe present invention. Particularly, FIG. 26 illustrates an example ofthe symbol interleaver when B=2112 and L=4096. Herein, B indicates ablock length in symbols that are outputted for symbol interleaving fromthe convolution encoder 513. And, L represents a block length in symbolsthat are actually interleaved by the symbol interleaver 514. At thispoint, the block length in symbols B inputted to the symbol interleaver514 is equivalent to 4×SOBL. More specifically, since one symbol isconfigured of 2 bits, the value of B may be set to be equal to 4×SOBL.

In the present invention, when performing the symbol-intereleavingprocess, the conditions of L=2^(m) (wherein m is an integer) and of L≧Bshould be satisfied. If there is a difference in value between B and L,(L−B) number of null (or dummy) symbols is added, thereby creating aninterleaving pattern, as shown in P′(i) of FIG. 26. Therefore, B becomesa block size of the actual symbols that are inputted to the symbolinterleaver 514 in order to be interleaved. L becomes an interleavingunit when the interleaving process is performed by an interleavingpattern created from the symbol interleaver 514.

Equation 6 shown below describes the process of sequentially receiving Bnumber of symbols, the order of which is to be rearranged, and obtainingan L value satisfying the conditions of L=2^(m) (wherein m is aninteger) and of L≧B, thereby creating the interleaving so as to realign(or rearrange) the symbol order.

In relation to all places, wherein 0≦i≦B−1,

P′(i)={89×i×(i+1)/2} mod L  Equation 6

Herein, L≧B, L=2^(m), wherein m is an integer.

As shown in P′(i) of FIG. 26, the order of B number of input symbols and(L-B) number of null symbols is rearranged by using the above-mentionedEquation 6. Then, as shown in P(i) of FIG. 26, the null byte places areremoved, so as to rearrange the order. Starting with the lowest value ofi, the P(i) are shifted to the left in order to fill the empty entrylocations. Thereafter, the symbols of the aligned interleaving patternP(i) are outputted to the symbol-byte converter 515 in order. Herein,the symbol-byte converter 515 converts to bytes the mobile service datasymbols, having the rearranging of the symbol order completed and thenoutputted in accordance with the rearranged order, and thereafteroutputs the converted bytes to the SCCC block-MPH block converter 516.The SCCC block-MPH block converter 516 converts the symbol-interleavedSCCC blocks to MPH blocks, which are then outputted to the groupformatter 303.

If the SCCC block mode value is equal to ‘00’, the SCCC block is mappedat a one-to-one (1:1) correspondence with each MPH block within the datagroup. In another example, if the SCCC block mode value is equal to‘01’, each SCCC block is mapped with two MPH blocks within the datagroup. For example, the SCCC block SCB1 is mapped with (B1, B6), theSCCC block SCB2 is mapped with (B2, B7), the SCCC block SCB3 is mappedwith (B3, B8), the SCCC block SCB4 is mapped with (B4, B9), and the SCCCblock SCB5 is mapped with (B5, B10). The MPH block that is outputtedfrom the SCCC block-MPH block converter 516 is configured of mobileservice data and FEC redundancy. In the present invention, the mobileservice data as well as the FEC redundancy of the MPH block will becollectively considered as mobile service data.

Group Formatter

The group formatter 303 inserts data of MPH blocks outputted from theblock processor 302 to the corresponding MPH blocks within the datagroup, which is formed in accordance with a pre-defined rule. Also, inassociation with the data-deinterleaving process, the group formatter303 inserts various place holders (or known data place holders) in thecorresponding region within the data group. More specifically, apartfrom the encoded mobile service data outputted from the block processor302, the group formatter 303 also inserts MPEG header place holders,non-systematic RS parity place holders, main service data place holders,which are associated with the data deinterleaving in a later process, asshown in FIG. 5.

Herein, the main service data place holders are inserted because themobile service data bytes and the main service data bytes arealternately mixed with one another in regions B to D based upon theinput of the data deinterleaver, as shown in FIG. 5. For example, basedupon the data outputted after data deinterleaving, the place holder forthe MPEG header may be allocated at the very beginning of each packet.Also, in order to configure an intended group format, dummy bytes mayalso be inserted. Furthermore, the group formatter 303 inserts placeholders for initializing the trellis encoding module 256 in thecorresponding regions. For example, the initialization data placeholders may be inserted in the beginning of the known data sequence.Additionally, the group formatter 303 may also insert signalinginformation, which are encoded and outputted from the signaling encoder304, in corresponding regions within the data group. At this point,reference may be made to the signaling information when the groupformatter 303 inserts each data type and respective place holders in thedata group. The process of encoding the signaling information andinserting the encoded signaling information to the data group will bedescribed in detail in a later process.

After inserting each data type and respective place holders in the datagroup, the group formatter 303 may deinterleave the data and respectiveplace holders, which have been inserted in the data group, as an inverseprocess of the data interleaver, thereby outputting the deinterleaveddata and respective place holders to the packet formatter 305. Morespecifically, when the data and respective place holders within the datagroup, which is configured (or structured) as shown in FIG. 5, aredeinterleaved by the group formatter 303 and outputted to the packetformatter 305, the structure of the data group may be identical to thestructure shown in FIG. 7. In order to do so, the group formatter 303may include a group format organizer 527, and a data deinterleaver 529,as shown in FIG. 27. The group format organizer 527 inserts data andrespective place holders in the corresponding regions within the datagroup, as described above. And, the data deinterleaver 529 deinterleavesthe inserted data and respective place holders as an inverse process ofthe data interleaver.

The packet formatter 305 removes the main service data place holders andthe RS parity place holders that were allocated for the deinterleavingprocess from the deinterleaved data being inputted. Then, the packetformatter 305 groups the remaining portion and inserts the 3-byte MPEGheader place holder in an MPEG header having a null packet PID (or anunused PID from the main service data packet). Furthermore, the packetformatter 305 adds a synchronization data byte at the beginning of each187-byte data packet. Also, when the group formatter 303 inserts knowndata place holders, the packet formatter 303 may insert actual knowndata in the known data place holders, or may directly output the knowndata place holders without any modification in order to make replacementinsertion in a later process. Thereafter, the packet formatter 305identifies the data within the packet-formatted data group, as describedabove, as a 188-byte unit mobile service data packet (i.e., MPEG TSpacket), which is then provided to the packet multiplexer 240.

Based upon the control of the control unit 200, the packet multiplexer240 multiplexes the data group packet-formatted and outputted from thepacket formatter 306 and the main service data packet outputted from thepacket jitter mitigator 220. Then, the packet multiplexer 240 outputsthe multiplexed data packets to the data randomizer 251 of thepost-processor 250. More specifically, the control unit 200 controls thetime-multiplexing of the packet multiplexer 240. If the packetmultiplexer 240 receives 118 mobile service data packets from the packetformatter 305, 37 mobile service data packets are placed before a placefor inserting VSB field synchronization. Then, the remaining 81 mobileservice data packets are placed after the place for inserting VSB fieldsynchronization. The multiplexing method may be adjusted by diversevariables of the system design. The multiplexing method and multiplexingrule of the packet multiplexer 240 will be described in more detail in alater process.

Also, since a data group including mobile service data in-between thedata bytes of the main service data is multiplexed (or allocated) duringthe packet multiplexing process, the shifting of the chronologicalposition (or place) of the main service data packet becomes relative.Also, a system object decoder (i.e., MPEG decoder) for processing themain service data of the receiving system, receives and decodes only themain service data and recognizes the mobile service data packet as anull data packet.

Therefore, when the system object decoder of the receiving systemreceives a main service data packet that is multiplexed with the datagroup, a packet jitter occurs.

At this point, since a multiple-level buffer for the video data existsin the system object decoder and the size of the buffer is relativelylarge, the packet jitter generated from the packet multiplexer 240 doesnot cause any serious problem in case of the video data. However, sincethe size of the buffer for the audio data in the object decoder isrelatively small, the packet jitter may cause considerable problem. Morespecifically, due to the packet jitter, an overflow or underflow mayoccur in the buffer for the main service data of the receiving system(e.g., the buffer for the audio data). Therefore, the packet jittermitigator 220 re-adjusts the relative position of the main service datapacket so that the overflow or underflow does not occur in the systemobject decoder.

In the present invention, examples of repositioning places for the audiodata packets within the main service data in order to minimize theinfluence on the operations of the audio buffer will be described indetail. The packet jitter mitigator 220 repositions the audio datapackets in the main service data section so that the audio data packetsof the main service data can be as equally and uniformly aligned andpositioned as possible. Additionally, when the positions of the mainservice data packets are relatively re-adjusted, associated programclock reference (PCR) values may also be modified accordingly. The PCRvalue corresponds to a time reference value for synchronizing the timeof the MPEG decoder. Herein, the PCR value is inserted in a specificregion of a TS packet and then transmitted.

In the example of the present invention, the packet jitter mitigator 220also performs the operation of modifying the PCR value. The output ofthe packet jitter mitigator 220 is inputted to the packet multiplexer240. As described above, the packet multiplexer 240 multiplexes the mainservice data packet outputted from the packet jitter mitigator 220 withthe mobile service data packet outputted from the pre-processor 230 intoa burst structure in accordance with a pre-determined multiplexing rule.Then, the packet multiplexer 240 outputs the multiplexed data packets tothe data randomizer 251 of the post-processor 250.

If the inputted data correspond to the main service data packet, thedata randomizer 251 performs the same randomizing process as that of theconventional randomizer. More specifically, the synchronization bytewithin the main service data packet is deleted. Then, the remaining 187data bytes are randomized by using a pseudo random byte generated fromthe data randomizer 251. Thereafter, the randomized data are outputtedto the RS encoder/non-systematic RS encoder 252.

On the other hand, if the inputted data correspond to the mobile servicedata packet, the data randomizer 251 may randomize only a portion of thedata packet. For example, if it is assumed that a randomizing processhas already been performed in advance on the mobile service data packetby the pre-processor 230, the data randomizer 251 deletes thesynchronization byte from the 4-byte MPEG header included in the mobileservice data packet and, then, performs the randomizing process only onthe remaining 3 data bytes of the MPEG header. Thereafter, therandomized data bytes are outputted to the RS encoder/non-systematic RSencoder 252. More specifically, the randomizing process is not performedon the remaining portion of the mobile service data excluding the MPEGheader. In other words, the remaining portion of the mobile service datapacket is directly outputted to the RS encoder/non-systematic RS encoder252 without being randomized. Also, the data randomizer 251 may or maynot perform a randomizing process on the known data (or known data placeholders) and the initialization data place holders included in themobile service data packet.

The RS encoder/non-systematic RS encoder 252 performs an RS encodingprocess on the data being randomized by the data randomizer 251 or onthe data bypassing the data randomizer 251, so as to add 20 bytes of RSparity data. Thereafter, the processed data are outputted to the datainterleaver 253. Herein, if the inputted data correspond to the mainservice data packet, the RS encoder/non-systematic RS encoder 252performs the same systematic RS encoding process as that of theconventional broadcasting system, thereby adding the 20-byte RS paritydata at the end of the 187-byte data. Alternatively, if the inputteddata correspond to the mobile service data packet, the RSencoder/non-systematic RS encoder 252 performs a non-systematic RSencoding process. At this point, the 20-byte RS parity data obtainedfrom the non-systematic RS encoding process are inserted in apre-decided parity byte place within the mobile service data packet.

The data interleaver 253 corresponds to a byte unit convolutionalinterleaver. The output of the data interleaver 253 is inputted to theparity replacer 254 and to the non-systematic RS encoder 255. Meanwhile,a process of initializing a memory within the trellis encoding module256 is primarily required in order to decide the output data of thetrellis encoding module 256, which is located after the parity replacer254, as the known data pre-defined according to an agreement between thereceiving system and the transmitting system. More specifically, thememory of the trellis encoding module 256 should first be initializedbefore the received known data sequence is trellis-encoded. At thispoint, the beginning portion of the known data sequence that is receivedcorresponds to the initialization data place holder and not to theactual known data. Herein, the initialization data place holder has beenincluded in the data by the group formatter within the pre-processor 230in an earlier process. Therefore, the process of generatinginitialization data and replacing the initialization data place holderof the corresponding memory with the generated initialization data arerequired to be performed immediately before the inputted known datasequence is trellis-encoded.

Additionally, a value of the trellis memory initialization data isdecided and generated based upon a memory status of the trellis encodingmodule 256. Further, due to the newly replaced initialization data, aprocess of newly calculating the RS parity and replacing the RS parity,which is outputted from the data interleaver 253, with the newlycalculated RS parity is required. Therefore, the non-systematic RSencoder 255 receives the mobile service data packet including theinitialization data place holders, which are to be replaced with theactual initialization data, from the data interleaver 253 and alsoreceives the initialization data from the trellis encoding module 256.

Among the inputted mobile service data packet, the initialization dataplace holders are replaced with the initialization data, and the RSparity data that are added to the mobile service data packet are removedand processed with non-systematic RS encoding. Thereafter, the new RSparity obtained by performing the non-systematic RS encoding process isoutputted to the parity replacer 255. Accordingly, the parity replacer255 selects the output of the data interleaver 253 as the data withinthe mobile service data packet, and the parity replacer 255 selects theoutput of the non-systematic RS encoder 255 as the RS parity. Theselected data are then outputted to the trellis encoding module 256.

Meanwhile, if the main service data packet is inputted or if the mobileservice data packet, which does not include any initialization dataplace holders that are to be replaced, is inputted, the parity replacer254 selects the data and RS parity that are outputted from the datainterleaver 253. Then, the parity replacer 254 directly outputs theselected data to the trellis encoding module 256 without anymodification. The trellis encoding module 256 converts the byte-unitdata to symbol units and performs a 12-way interleaving process so as totrellis-encode the received data. Thereafter, the processed data areoutputted to the synchronization multiplexer 260.

FIG. 28 illustrates a detailed diagram of one of 12 trellis encodersincluded in the trellis encoding module 256. Herein, the trellis encoderincludes first and second multiplexers 531 and 541, first and secondadders 532 and 542, and first to third memories 533, 542, and 544. Morespecifically, the first to third memories 533, 542, and 544 areinitialized by a set of trellis initialization data inserted in aninitialization data place holder by the parity replacer 254 and, then,outputted. More specifically, when the first two 2-bit symbols, whichare converted from each trellis initialization data byte, are inputted,the input bits of the trellis encoder will be replaced by the memoryvalues of the trellis encoder, as shown in FIG. 28.

Since 2 symbols (i.e., 4 bits) are required for trellis initialization,the last 2 symbols (i.e., 4 bits) from the trellis initialization bytesare not used for trellis initialization and are considered as a symbolfrom a known data byte and processed accordingly. When the trellisencoder is in the initialization mode, the input comes from an internaltrellis status (or state) and not from the parity replacer 254. When thetrellis encoder is in the normal mode, the input symbol provided fromthe parity replacer 254 will be processed. The trellis encoder providesthe converted (or modified) input data for trellis initialization to thenon-systematic RS encoder 255.

More specifically, when a selection signal designates a normal mode, thefirst multiplexer 531 selects an upper bit X2 of the input symbol. And,when a selection signal designates an initialization mode, the firstmultiplexer 531 selects the output of the first memory 533 and outputsthe selected output data to the first adder 532. The first adder 532adds the output of the first multiplexer 531 and the output of the firstmemory 533, thereby outputting the added result to the first memory 533and, at the same time, as a most significant (or uppermost) bit Z2. Thefirst memory 533 delays the output data of the first adder 532 by 1clock, thereby outputting the delayed data to the first multiplexer 531and the first adder 532.

Meanwhile, when a selection signal designates a normal mode, the secondmultiplexer 541 selects a lower bit X1 of the input symbol. And, when aselection signal designates an initialization mode, the secondmultiplexer 541 selects the output of the second memory 542, therebyoutputting the selected result to the second adder 543 and, at the sametime, as a lower bit Z1. The second adder 543 adds the output of thesecond multiplexer 541 and the output of the second memory 542, therebyoutputting the added result to the third memory 544. The third memory544 delays the output data of the second adder 543 by 1 clock, therebyoutputting the delayed data to the second memory 542 and, at the sametime, as a least significant (or lowermost) bit Z0. The second memory542 delays the output data of the third memory 544 by 1 clock, therebyoutputting the delayed data to the second adder 543 and the secondmultiplexer 541.

The synchronization multiplexer 260 inserts a field synchronizationsignal and a segment synchronization signal to the data outputted fromthe trellis encoding module 256 and, then, outputs the processed data tothe pilot inserter 271 of the transmission unit 270. Herein, the datahaving a pilot inserted therein by the pilot inserter 271 are modulatedby the modulator 272 in accordance with a pre-determined modulatingmethod (e.g., a VSB method). Thereafter, the modulated data aretransmitted to each receiving system though the radio frequency (RF)up-converter 273.

Multiplexing Method of Packet Multiplexer 240

Data of the error correction encoded and 1/H-rate encoded primary RSframe (i.e., when the RS frame mode value is equal to ‘00’) orprimary/secondary RS frame (i.e., when the RS frame mode value is equalto ‘01’), are divided into a plurality of data groups by the groupformatter 303. Then, the divided data portions are assigned to at leastone of regions A to D of each data group or to an MPH block among theMPH blocks B1 to B10, thereby being deinterleaved. Then, thedeinterleaved data group passes through the packet formatter 305,thereby being multiplexed with the main service data by the packetmultiplexer 240 based upon a de-decided multiplexing rule. The packetmultiplexer 240 multiplexes a plurality of consecutive data groups, sothat the data groups are assigned to be spaced as far apart from oneanother as possible within the sub-frame. For example, when it isassumed that 3 data groups are assigned to a sub-frame, the data groupsare assigned to a 1^(st) slot (Slot #0), a 5^(th) slot (Slot #4), and a9^(th) slot (Slot #8) in the sub-frame, respectively.

As described-above, in the assignment of the plurality of consecutivedata groups, a plurality of parades are multiplexed and outputted so asto be spaced as far apart from one another as possible within a sub-MPHframe. For example, the method of assigning data groups and the methodof assigning parades may be identically applied to all sub-frames foreach MPH frame or differently applied to each MPH frame.

FIG. 10 illustrates an example of a plurality of data groups included ina single parade, wherein the number of data groups included in asub-frame is equal to ‘3’, and wherein the data groups are assigned toan MPH frame by the packet multiplexer 240. Referring to FIG. 10, 3 datagroups are sequentially assigned to a sub-frame at a cycle period of 4slots. Accordingly, when this process is equally performed in the 5sub-frames included in the corresponding MPH frame, 15 data groups areassigned to a single MPH frame. Herein, the 15 data groups correspond todata groups included in a parade.

When data groups of a parade are assigned as shown in FIG. 10, thepacket multiplexer 240 may either assign main service data to each datagroup, or assign data groups corresponding to different parades betweeneach data group. More specifically, the packet multiplexer 240 mayassign data groups corresponding to multiple parades to one MPH frame.Basically, the method of assigning data groups corresponding to multipleparades is very similar to the method of assigning data groupscorresponding to a single parade. In other words, the packet multiplexer240 may assign data groups included in other parades to an MPH frameaccording to a cycle period of 4 slots. At this point, data groups of adifferent parade may be sequentially assigned to the respective slots ina circular method. Herein, the data groups are assigned to slotsstarting from the ones to which data groups of the previous parade havenot yet been assigned. For example, when it is assumed that data groupscorresponding to a parade are assigned as shown in FIG. 10, data groupscorresponding to the next parade may be assigned to a sub-frame startingeither from the 12^(th) slot of a sub-frame.

FIG. 11 illustrates an example of assigning and transmitting 3 parades(Parade #0, Parade #1, and Parade #2) to an MPH frame. For example, whenthe 1^(st) parade (Parade #0) includes 3 data groups for each sub-frame,the packet multiplexer 240 may obtain the positions of each data groupswithin the sub-frames by substituting values ‘0’ to ‘2’ for i inEquation 1. More specifically, the data groups of the 1^(st) parade(Parade #0) are sequentially assigned to the 1^(st), 5^(th), and 9^(th)slots (Slot #0, Slot #4, and Slot #8) within the sub-frame. Also, whenthe 2^(nd) parade includes 2 data groups for each sub-frame, the packetmultiplexer 240 may obtain the positions of each data groups within thesub-frames by substituting values ‘3’ and ‘4’ for i in Equation 1. Morespecifically, the data groups of the 2^(nd) parade (Parade #1) aresequentially assigned to the 2^(nd) and 12^(th) slots (Slot #3 and Slot#11) within the sub-frame. Finally, when the 3^(rd) parade includes 2data groups for each sub-frame, the packet multiplexer 240 may obtainthe positions of each data groups within the sub-frames by substitutingvalues ‘5’ and ‘6’ for i in Equation 1. More specifically, the datagroups of the 3^(rd) parade (Parade #2) are sequentially assigned andoutputted to the 7^(th) and 11^(th) slots (Slot #6 and Slot #10) withinthe sub-frame.

As described above, the packet multiplexer 240 may multiplex and outputdata groups of multiple parades to a single MPH frame, and, in eachsub-frame, the multiplexing process of the data groups may be performedserially with a group space of 4 slots from left to right. Therefore, anumber of groups of one parade per sub-frame (NOG) may correspond to anyone integer from ‘1’ to ‘8’. Herein, since one MPH frame includes 5sub-frames, the total number of data groups within a parade that can beallocated to an MPH frame may correspond to any one multiple of ‘5’ranging from ‘5’ to ‘40’.

Processing Signaling Information

The present invention assigns signaling information areas for insertingsignaling information to some areas within each data group. FIG. 29illustrates an example of assigning signaling information areas forinserting signaling information starting from the 1^(st) segment of the4^(th) MPH block (B4) to a portion of the 2^(nd) segment. Morespecifically, 276(=207+69) bytes of the 4^(th) MPH block (B4) in eachdata group are assigned as the signaling information area. In otherwords, the signaling information area consists of 207 bytes of the1^(st) segment and the first 69 bytes of the 2^(nd) segment of the4^(th) MPH block (B4). For example, the 1^(st) segment of the 4^(th) MPHblock (B4) corresponds to the 17^(th) or 173^(rd) segment of a VSBfield. The signaling information that is to be inserted in the signalinginformation area is FEC-encoded by the signaling encoder 304, therebyinputted to the group formatter 303.

The group formatter 303 inserts the signaling information, which isFEC-encoded and outputted by the signaling encoder 304, in the signalinginformation area within the data group. Herein, the signalinginformation may be identified by two different types of signalingchannels: a transmission parameter channel (TPC) and a fast informationchannel (FIC). Herein, the TPC information corresponds to signalinginformation including transmission parameters, such as RSframe-associated information, SCCC-associated information, and MPHframe-associated information. However, the signaling informationpresented herein is merely exemplary. And, since the adding or deletingof signaling information included in the TPC may be easily adjusted andmodified by one skilled in the art, the present invention will,therefore, not be limited to the examples set forth herein. Furthermore,the FIC is provided to enable a fast service acquisition of datareceivers, and the FIC includes cross layer information between thephysical layer and the upper layer(s).

FIG. 30 illustrates a detailed block diagram of the signaling encoder304 according to the present invention. Referring to FIG. 30, thesignaling encoder 304 includes a TPC encoder 561, an FIC encoder 562, ablock interleaver 563, a multiplexer 564, a signaling randomizer 565,and a PCCC encoder 566. The TPC encoder 561 receives 10-bytes of TPCdata and performs (18,10)-RS encoding on the 10-bytes of TPC data,thereby adding 8 bytes of parity data to the 10 bytes of TPC data. The18 bytes of RS-encoded TPC data are outputted to the multiplexer 564.The FIC encoder 562 receives 37-bytes of FIC data and performs(51,37)-RS encoding on the 37-bytes of FIC data, thereby adding 14 bytesof parity data to the 37 bytes of FIC data. Thereafter, the 51 bytes ofRS-encoded FIC data are inputted to the block interleaver 563, therebybeing interleaved in predetermined block units.

Herein, the block interleaver 563 corresponds to a variable length blockinterleaver. The block interleaver 563 interleaves the FIC data withineach sub-frame in TNoG(column)×51(row) block units and then outputs theinterleaved data to the multiplexer 564. Herein, the TNoG corresponds tothe total number of data groups being assigned to all sub-frames withinan MPH frame. The block interleaver 563 is synchronized with the firstset of FIC data in each sub-frame. The block interleaver 563 writes 51bytes of incoming (or inputted) RS codewords in a row direction (i.e.,row-by-row) and left-to-right and up-to-down directions and reads 51bytes of RS codewords in a column direction (i.e., column-by-column) andleft-to-right and up-to-down directions, thereby outputting the RScodewords.

The multiplexer 564 multiplexes the RS-encoded TPC data from the TPCencoder 561 and the block-interleaved FIC data from the blockinterleaver 563 along a time axis. Then, the multiplexer 564 outputs 69bytes of the multiplexed data to the signaling randomizer 565. Thesignaling randomizer 565 randomizes the multiplexed data and outputs therandomized data to the PCCC encoder 566. The signaling randomizer 565may use the same generator polynomial of the randomizer used for mobileservice data. Also, initialization occurs in each data group. The PCCCencoder 566 corresponds to an inner encoder performing PCCC-encoding onthe randomized data (i.e., signaling information data). The PCCC encoder566 may include 6 even component encoders and 6 odd component encoders.

FIG. 31 illustrates an example of a syntax structure of TPC data beinginputted to the TPC encoder 561. The TPC data are inserted in thesignaling information area of each data group and then transmitted. TheTPC data may include a sub-frame_number field, a slot_number field, aparade_id field, a starting_group_number (SGN) field, a number_of_groups(NoG) field, a parade_repetition_cycle (PRC) field, an RS_frame_modefield, an RS_code_mode_primary field, an RS_code_mode_secondary field,an SCCC_block_mode field, an SCCC_outer_code_mode_A field, anSCCC_outer_code_mode_B field, an SCCC_outer_code_mode_C field, anSCCC_outer_code_mode_D field, an FIC_version field, aparade_continuity_counter field, and a TNoG field.

The Sub-Frame_number field corresponds to the current Sub-Frame numberwithin the MPH frame, which is transmitted for MPH framesynchronization. The value of the Sub-Frame_number field may range from0 to 4. The Slot_number field indicates the current slot number withinthe sub-frame, which is transmitted for MPH frame synchronization. Also,the value of the Sub-Frame_number field may range from 0 to 15. TheParade_id field identifies the parade to which this group belongs. Thevalue of this field may be any 7-bit value. Each parade in a MPHtransmission shall have a unique Parade_id field.

Communication of the Parade_id between the physical layer and themanagement layer may be performed by means of an Ensemble_id fieldformed by adding one bit to the left of the Parade_id field. If theEnsemble_id field is used for the primary Ensemble delivered throughthis parade, the added MSB shall be equal to ‘0’. Otherwise, if theEnsemble_id field is used for the secondary ensemble, the added MSBshall be equal to ‘1’. Assignment of the Parade_id field values mayoccur at a convenient level of the system, usually in the managementlayer. The starting_group_number (SGN) field shall be the firstSlot_number for a parade to which this group belongs, as determined byEquation 1 (i.e., after the Slot numbers for all preceding parades havebeen calculated). The SGN and NoG shall be used according to Equation 1to obtain the slot numbers to be allocated to a parade within thesub-frame.

The number_of_Groups (NoG) field shall be the number of groups in asub-frame assigned to the parade to which this group belongs, minus 1,e.g., NoG=0 implies that one group is allocated (or assigned) to thisparade in a sub-frame. The value of NoG may range from 0 to 7. Thislimits the amount of data that a parade may take from the main (legacy)service data, and consequently the maximum data that can be carried byone parade. The slot numbers assigned to the corresponding Parade can becalculated from SGN and NoG, using Equation 1. By taking each parade insequence, the specific slots for each parade will be determined, andconsequently the SGN for each succeeding parade. For example, if for aspecific parade SGN=3 and NoG=3 (010b for 3-bit field of NoG),substituting i=3, 4, and 5 in Equation 1 provides slot numbers 12, 2,and 6. The Parade_repetition_cycle (PRC) field corresponds to the cycletime over which the parade is transmitted, minus 1, specified in unitsof MPH frames, as described in Table 12.

TABLE 12 PRC Description 000 This parade shall be transmitted once everyMPH frame. 001 This parade shall be transmitted once every 2 MPH frames.010 This parade shall be transmitted once every 3 MPH frames. 011 Thisparade shall be transmitted once every 4 MPH frames. 100 This paradeshall be transmitted once every 5 MPH frames. 101 This parade shall betransmitted once every 6 MPH frames. 110 This parade shall betransmitted once every 7 MPH frames. 111 Reserved

The RS_Frame_mode field shall be as defined in Table 1. TheRS_code_mode_primary field shall be the RS code mode for the primary RSframe. Herein, the RS code mode is defined in Table 6. TheRS_code_mode_secondary field shall be the RS code mode for the secondaryRS frame. Herein, the RS code mode is defined in Table 6. TheSCCC_Block_mode field shall be as defined in Table 7. TheSCCC_outer_code_mode_A field corresponds to the SCCC outer code mode forRegion A. The SCCC outer code mode is defined in Table 8. TheSCCC_outer_code_mode_B field corresponds to the SCCC outer code mode forRegion B. The SCCC_outer_code_mode_C field corresponds be the SCCC outercode mode for Region C. And, the SCCC_outer_code_mode_D fieldcorresponds to the SCCC outer code mode for Region D.

The FIC_version field may be supplied by the management layer (whichalso supplies the FIC data). The Parade_continuity_counter field countermay increase from 0 to 15 and then repeat its cycle. This counter shallincrement by 1 every (PRC+1) MPH frames. For example, as shown in Table12, PRC=011 (decimal 3) implies that Parade_continuity_counter increasesevery fourth MPH frame. The TNOG field may be identical for allsub-frames in an MPH Frame. However, the information included in the TPCdata presented herein is merely exemplary. And, since the adding ordeleting of information included in the TPC may be easily adjusted andmodified by one skilled in the art, the present invention will,therefore, not be limited to the examples set forth herein.

Since the TPC parameters (excluding the Sub-Frame_number field and theSlot_number field) for each parade do not change their values during anMPH frame, the same information is repeatedly transmitted through allMPH groups belonging to the corresponding parade during an MPH frame.This allows very robust and reliable reception of the TPC data. Becausethe Sub-Frame_number and the Slot_number are increasing counter values,they also are robust due to the transmission of regularly expectedvalues.

Furthermore, the FIC information is provided to enable a fast serviceacquisition of data receivers, and the FIC information includes crosslayer information between the physical layer and the upper layer(s).

FIG. 32 illustrates an example of a transmission scenario of the TPCdata and the FIC data. The values of the Sub-Frame_number field,Slot_number field, Parade_id field, Parade_repetition_cycle field, andParade_continuity_counter field may corresponds to the current MPH framethroughout the 5 sub-frames within a specific MPH frame. Some of TPCparameters and FIC data are signaled in advance. The SGN, NoG and allFEC modes may have values corresponding to the current MPH frame in thefirst two sub-frames. The SGN, NoG and all FEC modes may have valuescorresponding to the frame in which the parade next appears throughoutthe 3^(rd), 4^(th) and 5^(th) sub-frames of the current MPH frame. Thisenables the MPH receivers to receive (or acquire) the transmissionparameters in advance very reliably.

For example, when Parade_repetition_cycle=‘000’, the values of the3^(rd), 4^(th), and 5^(th) sub-frames of the current MPH framecorrespond to the next MPH frame. Also, whenParade_repetition_cycle=‘011’, the values of the 3^(rd), 4^(th), and5^(th) sub-frames of the current MPH frame correspond to the 4^(th) MPHframe and beyond. The FIC_version field and the FIC_data field may havevalues that apply to the current MPH Frame during the 1^(st) sub-frameand the 2^(nd) sub-frame, and they shall have values corresponding tothe MPH frame immediately following the current MPH frame during the3^(rd), 4^(th), and 5^(th) sub-frames of the current MPH frame.

Meanwhile, the receiving system may turn the power on only during a slotto which the data group of the designated (or desired) parade isassigned, and the receiving system may turn the power off during theremaining slots, thereby reducing power consumption of the receivingsystem. Such characteristic is particularly useful in portable or mobilereceivers, which require low power consumption. For example, it isassumed that data groups of a 1^(st) parade with NOG=3, a 2^(nd) paradewith NOG=2, and a 3^(rd) parade with NOG=3 are assigned to one MPHframe, as shown in FIG. 33. It is also assumed that the user hasselected a mobile service included in the 1^(st) parade using the keypadprovided on the remote controller or terminal. In this case, thereceiving system turns the power on only during a slot that data groupsof the 1^(st) parade is assigned, as shown in FIG. 33, and turns thepower off during the remaining slots, thereby reducing powerconsumption, as described above. At this point, the power is required tobe turned on briefly earlier than the slot to which the actualdesignated data group is assigned (or allocated). This is to enable thetuner or demodulator to converge in advance.

Assignment of Known Data (or Training Signal)

In addition to the payload data, the MPH transmission system insertslong and regularly spaced training sequences into each group. Theregularity is an especially useful feature since it provides thegreatest possible benefit for a given number of training symbols inhigh-Doppler rate conditions. The length of the training sequences isalso chosen to allow fast acquisition of the channel during burstedpower-saving operation of the demodulator. Each group contains 6training sequences. The training sequences are specified beforetrellis-encoding. The training sequences are then trellis-encoded andthese trellis-encoded sequences also are known sequences. This isbecause the trellis encoder memories are initialized to pre-determinedvalues at the beginning of each sequence. The form of the 6 trainingsequences at the byte level (before trellis-encoding) is shown in FIG.34. This is the arrangement of the training sequence at the groupformatter 303.

The 1^(st) training sequence is located at the last 2 segments of the3^(rd) MPH block (B3). The 2^(nd) training sequence may be inserted atthe 2^(nd) and 3^(rd) segments of the 4^(th) MPH block (B4). The 2^(nd)training sequence is next to the signaling area, as shown in FIG. 5.Then, the 3^(rd) training sequence, the 4^(th) training sequence, the5^(th) training sequence, and the 6^(th) training sequence may be placedat the last 2 segments of the 4^(th), 5^(th), 6^(th), and 7^(th) MPHblocks (B4, B5, B6, and B7), respectively. As shown in FIG. 34, the1^(st) training sequence, the 3^(rd) training sequence, the 4^(th)training sequence, the 5^(th) training sequence, and the 6^(th) trainingsequence are spaced 16 segments apart from one another. Referring toFIG. 34, the dotted area indicates trellis initialization data bytes,the lined area indicates training data bytes, and the white areaincludes other bytes such as the FEC-coded MPH service data bytes,FEC-coded signaling data, main service data bytes, RS parity data bytes(for backwards compatibility with legacy ATSC receivers) and/or dummydata bytes.

FIG. 35 illustrates the training sequences (at the symbol level) aftertrellis-encoding by the trellis encoder. Referring to FIG. 35, thedotted area indicates data segment sync symbols, the lined areaindicates training data symbols, and the white area includes othersymbols, such as FEC-coded mobile service data symbols, FEC-codedsignaling data, main service data symbols, RS parity data symbols (forbackwards compatibility with legacy ATSC receivers), dummy data symbols,trellis initialization data symbols, and/or the first part of thetraining sequence data symbols. Due to the intra-segment interleaving ofthe trellis encoder, various types of data symbols will be mixed in thewhite area.

After the trellis-encoding process, the last 1416 (=588+828) symbols ofthe 1^(st) training sequence, the 3^(rd) training sequence, the 4^(th)training sequence, the 5^(th) training sequence, and the 6^(th) trainingsequence commonly share the same data pattern. Including the datasegment synchronization symbols in the middle of and after eachsequence, the total length of each common training pattern is 1424symbols. The 2^(nd) training sequence has a first 528-symbol sequenceand a second 528-symbol sequence that have the same data pattern. Morespecifically, the 528-symbol sequence is repeated after the 4-symboldata segment synchronization signal. At the end of each trainingsequence, the memory contents of the twelve modified trellis encodersshall be set to zero(0).

Demodulating Unit within Receiving System

FIG. 36 illustrates an example of a demodulating unit in a digitalbroadcast receiving system according to the present invention. Thedemodulating unit of FIG. 36 uses known data information, which isinserted in the mobile service data section and, then, transmitted bythe transmitting system, so as to perform carrier synchronizationrecovery, frame synchronization recovery, and channel equalization,thereby enhancing the receiving performance. Also the demodulating unitmay turn the power on only during a slot to which the data group of thedesignated (or desired) parade is assigned, thereby reducing powerconsumption of the receiving system.

Referring to FIG. 36, the demodulating unit includes a demodulator 1002,an equalizer 1003, a known sequence detector 1004, a block decoder 1005,a RS frame decoder 1006, a derandomizer 1007. The demodulating unit mayfurther include a data deinterleaver 1009, a RS decoder 1010, and a dataderandomizer 1011. The demodulating unit may further include a signalinginformation decoder 1013. The receiving system also may further includea power controller 5000 for controlling power supply of the demodulatingunit.

Herein, for simplicity of the description of the present invention, theRS frame decoder 1006, and the derandomizer 1007 will be collectivelyreferred to as a mobile service data processing unit. And, the datadeinterleaver 1009, the RS decoder 1010, and the data derandomizer 1011will be collectively referred to as a main service data processing unit.More specifically, a frequency of a particular channel tuned by a tunerdown converts to an intermediate frequency (IF) signal. Then, thedown-converted data 1001 outputs the down-converted IF signal to thedemodulator 1002 and the known sequence detector 1004. At this point,the down-converted data 1001 is inputted to the demodulator 1002 and theknown sequence detector 1004 via analog/digital converter ADC (notshown). The ADC converts pass-band analog IF signal into pass-banddigital IF signal.

The demodulator 1002 performs self gain control, carrier recovery, andtiming recovery processes on the inputted pass-band digital IF signal,thereby modifying the IF signal to a base-band signal. Then, thedemodulator 1002 outputs the newly created base-band signal to theequalizer 1003 and the known sequence detector 1004. The equalizer 1003compensates the distortion of the channel included in the demodulatedsignal and then outputs the error-compensated signal to the blockdecoder 1005.

At this point, the known sequence detector 1004 detects the knownsequence place inserted by the transmitting end from the input/outputdata of the demodulator 1002 (i.e., the data prior to the demodulationprocess or the data after the demodulation process). Thereafter, theplace information along with the symbol sequence of the known data,which are generated from the detected place, is outputted to thedemodulator 1002 and the equalizer 1003. Also, the known data detector1004 outputs a set of information to the block decoder 1005. This set ofinformation is used to allow the block decoder 1005 of the receivingsystem to identify the mobile service data that are processed withadditional encoding from the transmitting system and the main servicedata that are not processed with additional encoding. In addition,although the connection status is not shown in FIG. 36, the informationdetected from the known data detector 1004 may be used throughout theentire receiving system and may also be used in the RS frame decoder1006.

The demodulator 1002 uses the known data symbol sequence during thetiming and/or carrier recovery, thereby enhancing the demodulatingperformance. Similarly, the equalizer 1003 uses the known data so as toenhance the equalizing performance. Moreover, the decoding result of theblock decoder 1005 may be fed-back to the equalizer 1003, therebyenhancing the equalizing performance.

Power On/Off Control

The data demodulated in the demodulator 1002 or the data equalized inthe channel equalizer 1003 is inputted to the signaling informationdecoder 1013. The known data information detected in the known sequencedetector 1004 is inputted to the signaling information decoder 1013. Thesignaling information decoder 1013 extracts and decodes signalinginformation from the inputted data, the decoded signaling informationprovides to blocks requiring the signaling information. For example, theSCCC-associated information may output to the block decoder 1005, andthe RS frame-associated information may output to the RS frame decoder1006. The MPH frame-associated information may output to the knownsequence detector 1004 and the power controller 5000.

Herein, the RS frame-associated information may include RS frame modeinformation and RS code mode information. The SCCC-associatedinformation may include SCCC block mode information and SCCC outer codemode information. The MPH frame-associated information may includesub-frame count information, slot count information, parade_idinformation, SGN information, NoG information, and so on, as shown inFIG. 32.

More specifically, the signaling information between first known dataarea and second known data area can know by using known data informationbeing outputted in the known sequence detector 1004. Therefore, thesignaling information decoder 1013 may extract and decode signalinginformation from the data being outputted in the demodulator 1002 or thechannel equalizer 1003.

The power controller 5000 is inputted the MPH frame-associatedinformation from the signaling information decoder 1013, and controlspower of the tuner and the demodulating unit.

According to the embodiment of the present invention, the powercontroller 5000 turns the power on only during a slot to which a slot ofthe parade including user-selected mobile service is assigned. The powercontroller 5000 then turns the power off during the remaining slots.

For example, it is assumed that data groups of a 1^(st) parade withNOG=3, a 2^(nd) parade with NOG=2, and a 3^(rd) parade with NOG=3 areassigned to one MPH frame, as shown in FIG. 33. It is also assumed thatthe user has selected a mobile service included in the 1^(st) paradeusing the keypad provided on the remote controller or terminal. In thiscase, the power controller 5000 turns the power on only during a slotthat data groups of the 1^(st) parade is assigned, as shown in FIG. 33,and turns the power off during the remaining slots, thereby reducingpower consumption.

Demodulator and Known Sequence Detector

At this point, the transmitting system may receive a data frame (or VSBframe) including a data group which known data sequence (or trainingsequence) is periodically inserted therein. Herein, the data group isdivided into regions A to D, as shown in FIG. 5. More specifically, inthe example of the present invention, each region A, B, C, and D arefurther divided into MPH blocks B4 to B7, MPH blocks B3 and B8, MPHblocks B2 and B9, MPH blocks B1 and B10, respectively.

FIG. 37 illustrates an example of known data sequence being periodicallyinserted and transmitted in-between actual data by the transmittingsystem. Referring to FIG. 37, AS represents the number of valid datasymbols, and BS represents the number of known data symbols. Therefore,BS number of known data symbols are inserted and transmitted at a periodof (AS+BS) symbols. Herein, AS may correspond to mobile service data,main service data, or a combination of mobile service data and mainservice data. In order to be differentiated from the known data, datacorresponding to AS will hereinafter be referred to as valid data.

Referring to FIG. 37, known data sequence having the same pattern areincluded in each known data section that is being periodically inserted.Herein, the length of the known data sequence having identical datapatterns may be either equal to or different from the length of theentire (or total) known data sequence of the corresponding known datasection (or block). If the two lengths are different from one another,the length of the entire known data sequence should be longer than thelength of the known data sequence having identical data patterns. Inthis case, the same known data sequences are included in the entireknown data sequence. The known sequence detector 1004 detects theposition of the known data being periodically inserted and transmittedas described above. At the same time, the known sequence detector 1004may also estimate initial frequency offset during the process ofdetecting known data. In this case, the demodulator 1002 may estimatewith more accuracy carrier frequency offset from the information on theknown data position (or known sequence position indicator) and initialfrequency offset estimation value, thereby compensating the estimatedinitial frequency offset.

FIG. 38 illustrates a detailed block diagram of a demodulator accordingto the present invention. Referring to FIG. 38, the demodulator includesa phase splitter 1010, a numerically controlled oscillator (NCO) 1020, afirst multiplier 1030, a resampler 1040, a second multiplier 1050, amatched filter 1060, a DC remover 1070, a timing recovery unit 1080, acarrier recovery unit 1090, and a phase compensator 1110. Herein, theknown sequence detector 1004 includes a known sequence detector andinitial frequency offset estimator 1004-1 for estimating known datainformation and initial frequency offset. Also referring to FIG. 38, thephase splitter 1010 receives a pass band digital signal and splits thereceived signal into a pass band digital signal of a real number elementand a pass band digital signal of an imaginary number element bothhaving a phase of 90 degrees between one another. In other words, thepass band digital signal is split into complex signals. The splitportions of the pass band digital signal are then outputted to the firstmultiplier 1030. Herein, the real number signal outputted from the phasesplitter 1010 will be referred to as an ‘I’ signal, and the imaginarynumber signal outputted from the phase splitter 1010 will be referred toas a ‘Q’ signal, for simplicity of the description of the presentinvention.

The first multiplier 1030 multiplies the I and Q pass band digitalsignals, which are outputted from the phase splitter 1010, to a complexsignal having a frequency proportional to a constant being outputtedfrom the NCO 1020, thereby changing the I and Q pass band digitalsignals to baseband digital complex signals. Then, the baseband digitalsignals of the first multiplier 1030 are inputted to the resampler 1040.The resampler 1040 resamples the signals being outputted from the firstmultiplier 1030 so that the signal corresponds to the timing clockprovided by the timing recovery unit 1080. Thereafter, the resampler1040 outputs the resampled signals to the second multiplier 1050.

For example, when the analog/digital converter uses a 25 MHz fixedoscillator, the baseband digital signal having a frequency of 25 MHz,which is created by passing through the analog/digital converter, thephase splitter 1010, and the first multiplier 1030, is processed with aninterpolation process by the resampler 1040. Thus, the interpolatedsignal is recovered to a baseband digital signal having a frequencytwice that of the receiving signal of a symbol clock (i.e., a frequencyof 21.524476 MHz). Alternatively, if the analog/digital converter usesthe timing clock of the timing recovery unit 1080 as the samplingfrequency (i.e., if the analog/digital converter uses a variablefrequency) in order to perform an A/D conversion process, the resampler1040 is not required and may be omitted.

The second multiplier 1050 multiplies an output frequency of the carrierrecovery unit 1090 with the output of the resampler 1040 so as tocompensate any remaining carrier included in the output signal of theresampler 1040. Thereafter, the compensated carrier is outputted to thematched filter 1060 and the timing recovery unit 1080. The signalmatched-filtered by the matched filter 1060 is inputted to the DCremover 1070, the known sequence detector and initial frequency offsetestimator 1004-1, and the carrier recovery unit 1090.

The known sequence detector and initial frequency offset estimator1004-1 detects the place (or position) of the known data sequences thatare being periodically or non-periodically transmitted. Simultaneously,the known sequence detector and initial frequency offset estimator1004-1 estimates an initial frequency offset during the known sequencedetection process. More specifically, while the transmission data frameis being received, as shown in FIG. 5, the known sequence detector andinitial frequency offset estimator 1004-1 detects the position (orplace) of the known data included in the transmission data frame. Then,the known sequence detector and initial frequency offset estimator1004-1 outputs the detected information on the known data place (i.e., aknown sequence position indicator) to the timing recovery unit 1080, thecarrier recovery unit 1090, and the phase compensator 1110 of thedemodulator 1002 and the equalizer 1003. Furthermore, the known sequencedetector and initial frequency offset estimator 1004-1 estimates theinitial frequency offset, which is then outputted to the carrierrecovery unit 1090. At this point, the known sequence detector andinitial frequency offset estimator 1004-1 may either receive the outputof the matched filter 1060 or receive the output of the resampler 1040.This may be optionally decided depending upon the design of the systemdesigner.

The timing recovery unit 1080 uses the output of the second multiplier1050 and the known sequence position indicator detected from the knownsequence detector and initial frequency offset estimator 1004-1, so asto detect the timing error and, then, to output a sampling clock beingin proportion with the detected timing error to the resampler 1040,thereby adjusting the sampling timing of the resampler 1040. At thispoint, the timing recovery unit 1080 may receive the output of thematched filter 1060 instead of the output of the second multiplier 1050.This may also be optionally decided depending upon the design of thesystem designer.

Meanwhile, the DC remover 1070 removes a pilot tone signal (i.e., DCsignal), which has been inserted by the transmitting system, from thematched-filtered signal. Thereafter, the DC remover 1070 outputs theprocessed signal to the phase compensator 1110. The phase compensator1110 uses the data having the DC removed by the DC remover 1070 and theknown sequence position indicator detected by the known sequencedetector and initial frequency offset estimator 1004-1 to estimate thefrequency offset and, then, to compensate the phase change included inthe output of the DC remover 1070. The data having its phase changecompensated are inputted to the equalizer 1003. Herein, the phasecompensator 1110 is optional. If the phase compensator 1110 is notprovided, then the output of the DC remover 1070 is inputted to theequalizer 1003 instead.

FIG. 39 includes detailed block diagrams of the timing recovery unit1080, the carrier recovery unit 1090, and the phase compensator 1110 ofthe demodulator. According to an embodiment of the present invention,the carrier recovery unit 1090 includes a buffer 1091, a frequencyoffset estimator 1092, a loop filter 1093, a holder 1094, an adder 1095,and a NCO 1096. Herein, a decimator may be included before the buffer1091. The timing recovery unit 1080 includes a decimator 1081, a buffer1082, a timing error detector 1083, a loop filter 1084, a holder 1085,and a NCO 1086. Finally, the phase compensator 1110 includes a buffer1111, a frequency offset estimator 1112, a holder 1113, a NCO 1114, anda multiplier 1115. Furthermore, a decimator 1200 may be included betweenthe phase compensator 1110 and the equalizer 1003. The decimator 1200may be outputted in front of the DC remover 1070 instead of at theoutputting end of the phase compensator 1110.

Herein, the decimators correspond to components required when a signalbeing inputted to the demodulator is oversampled to N times by theanalog/digital converter. More specifically, the integer N representsthe sampling rate of the received signal. For example, when the inputsignal is oversampled to 2 times (i.e., when N=2) by the analog/digitalconverter, this indicates that two samples are included in one symbol.In this case, each of the decimators corresponds to a ½ decimator.Depending upon whether or not the oversampling process of the receivedsignal has been performed, the signal may bypass the decimators.

Meanwhile, the output of the second multiplier 1050 is temporarilystored in the decimator 1081 and the buffer 1082 both included in thetiming recovery unit 1080. Subsequently, the temporarily stored outputdata are inputted to the timing error detector 1083 through thedecimator 1081 and the buffer 1082. Assuming that the output of thesecond multiplier 1050 is oversampled to N times its initial state, thedecimator 1081 decimates the output of the second multiplier 1050 at adecimation rate of 1/N. Then, the 1/N-decimated data are inputted to thebuffer 1082. In other words, the decimator 1081 performs decimation onthe input signal in accordance with a VSB symbol cycle. Furthermore, thedecimator 1081 may also receive the output of the matched filter 1060instead of the output of the second multiplier 1050. The timing errordetector 1083 uses the data prior to or after being processed withmatched-filtering and the known sequence position indicator outputtedfrom the known sequence detector and initial frequency offset estimator1004-1 in order to detect a timing error. Thereafter, the detectedtiming error is outputted to the loop filter 1084. Accordingly, thedetected timing error information is obtained once during eachrepetition cycle of the known data sequence.

For example, if a known data sequence having the same pattern isperiodically inserted and transmitted, as shown in FIG. 37, the timingerror detector 1083 may use the known data in order to detect the timingerror. There exists a plurality of methods for detecting timing error byusing the known data. In the example of the present invention, thetiming error may be detected by using a correlation characteristicbetween the known data and the received data in the time domain, theknown data being already known in accordance with a pre-arrangedagreement between the transmitting system and the receiving system. Thetiming error may also be detected by using the correlationcharacteristic of the two known data types being received in thefrequency domain. Thus, the detected timing error is outputted. Inanother example, a spectral lining method may be applied in order todetect the timing error. Herein, the spectral lining method correspondsto a method of detecting timing error by using sidebands of the spectrumincluded in the received signal.

The loop filter 1084 filters the timing error detected by the timingerror detector 1083 and, then, outputs the filtered timing error to theholder 1085. The holder 1085 holds (or maintains) the timing errorfiltered and outputted from the loop filter 1084 during a pre-determinedknown data sequence cycle period and outputs the processed timing errorto the NCO 1086. Herein, the order of positioning of the loop filter1084 and the holder 1085 may be switched with one another. Inadditionally, the function of the holder 1085 may be included in theloop filter 1084, and, accordingly, the holder 1085 may be omitted. TheNCO 1086 accumulates the timing error outputted from the holder 1085.Thereafter, the NCO 1086 outputs the phase element (i.e., a samplingclock) of the accumulated timing error to the resampler 1040, therebyadjusting the sampling timing of the resampler 1040.

Meanwhile, the buffer 1091 of the carrier recovery unit 1090 may receiveeither the data inputted to the matched filter 1060 or the dataoutputted from the matched filter 1060 and, then, temporarily store thereceived data. Thereafter, the temporarily stored data are outputted tothe frequency offset estimator 1092. If a decimator is provided in frontof the buffer 1091, the input data or output data of the matched filter1060 are decimated by the decimator at a decimation rate of 1/N.Thereafter, the decimated data are outputted to the buffer 1091. Forexample, when the input data or output data of the matched filter 1060are oversampled to 2 times (i.e., when N=2), this indicates that theinput data or output data of the matched filter 1060 are decimated at arate of ½ by the decimator 1081 and then outputted to the buffer 1091.More specifically, when a decimator is provided in front of the buffer1091, the carrier recovery unit 1090 operates in symbol units.Alternatively, if a decimator is not provided, the carrier recovery unit1090 operates in oversampling units.

The frequency offset estimator 1092 uses the input data or output dataof the matched filter 1060 and the known sequence position indicatoroutputted from the known sequence detector and initial frequency offsetestimator 1004-1 in order to estimate the frequency offset. Then, theestimated frequency offset is outputted to the loop filter 1093.Therefore, the estimated frequency offset value is obtained once everyrepetition period of the known data sequence. The loop filter 1093performs low pass filtering on the frequency offset value estimated bythe frequency offset estimator 1092 and outputs the low pass-filteredfrequency offset value to the holder 1094. The holder 1094 holds (ormaintains) the low pass-filtered frequency offset value during apre-determined known data sequence cycle period and outputs thefrequency offset value to the adder 1095. Herein, the positions of theloop filter 1093 and the holder 1094 may be switched from one to theother. Furthermore, the function of the holder 1085 may be included inthe loop filter 1093, and, accordingly, the holder 1094 may be omitted.

The adder 1095 adds the value of the initial frequency offset estimatedby the known sequence detector and initial frequency offset estimator1004-1 to the frequency offset value outputted from the loop filter 1093(or the holder 1094). Thereafter, the added offset value is outputted tothe NCO 1096. Herein, if the adder 1095 is designed to also receive theconstant being inputted to the NCO 1020, the NCO 1020 and the firstmultiplier 1030 may be omitted. In this case, the second multiplier 1050may simultaneously perform changing signals to baseband signals andremoving remaining carrier.

The NCO 1096 generates a complex signal corresponding to the frequencyoffset outputted from the adder 1095, which is then outputted to thesecond multiplier 1050. Herein, the NCO 1096 may include a ROM. In thiscase, the NCO 1096 generates a compensation frequency corresponding tothe frequency offset being outputted from the adder 1095. Then, the NCO1096 reads a complex cosine corresponding to the compensation frequencyfrom the ROM, which is then outputted to the second multiplier 1050. Thesecond multiplier 1050 multiplies the output of the NCO 1094 included inthe carrier recovery unit 1090 to the output of the resampler 1040, soas to remove the carrier offset included in the output signal of theresampler 1040.

FIG. 40 illustrates a detailed block diagram of the frequency offsetestimator 1092 of the carrier recovery unit 1090 according to anembodiment of the present invention. Herein, the frequency offsetestimator 1092 operates in accordance with the known sequence positionindicator detected from the known sequence detector and initialfrequency offset estimator 1004-1. At this point, if the input data oroutput data of the matched filter 1060 are inputted through thedecimator, the frequency offset estimator 1092 operates in symbol units.Alternatively, if a decimator is not provided, the frequency offsetestimator 1092 operates in oversampling units. In the example given inthe description of the present invention, the frequency offset estimator1092 operates in symbol units. Referring to FIG. 40, the frequencyoffset estimator 1092 includes a controller 1310, a first N symbolbuffer 1301, a K symbol delay 1302, a second N symbol buffer 1303, aconjugator 1304, a multiplier 1305, an accumulator 1306, a phasedetector 1307, a multiplier 1308, and a multiplexer 1309. The frequencyoffset estimator 1092 having the above-described structure, as shown inFIG. 40, will now be described in detail with respect to an operationexample during a known data section.

The first N symbol buffer 1301 may store a maximum of N number of symbolbeing inputted thereto. The symbol data that are temporarily stored inthe first N symbol buffer 1301 are then inputted to the multiplier 1305.At the same time, the inputted symbol is inputted to the K symbol delay1302 so as to be delayed by K symbols. Thereafter, the delayed symbolpasses through the second N symbol buffer 1303 so as to be conjugated bythe conjugator 1304. Thereafter, the conjugated symbol is inputted tothe multiplier 1305. The multiplier 1305 multiplies the output of thefirst N symbol buffer 1301 and the output of the conjugator 1304. Then,the multiplier 1305 outputs the multiplied result to the accumulator1306. Subsequently, the accumulator 1306 accumulates the output of themultiplier 1305 during N symbol periods, thereby outputted theaccumulated result to the phase detector 1307.

The phase detector 1307 extracts the corresponding phase informationfrom the output of the accumulator 1306, which is then outputted to themultiplier 1308. The multiplier 1308 then divides the phase informationby K, thereby outputting the divided result to the multiplexer 1309.Herein, the result of the phase information divided by becomes thefrequency offset estimation value. More specifically, at the point wherethe input of the known data ends or at a desired point, the frequencyoffset estimator 1092 accumulates during an N symbol periodmultiplication of the complex conjugate of N number of the input datastored in the first N symbol buffer 1301 and the complex conjugate ofthe N number of the input data that are delayed by K symbols and storedin the second N symbol buffer 1303. Thereafter, the accumulated value isdivided by K, thereby extracting the frequency offset estimation value.

Based upon a control signal of the controller 1310, the multiplexer 1309selects either the output of the multiplier 1308 or ‘0’ and, then,outputs the selected result as the final frequency offset estimationvalue. The controller 1310 receives the known data sequence positionindicator from the known sequence detector and initial frequency offsetestimator 1004-1 in order to control the output of the multiplexer 1309.More specifically, the controller 1310 determines based upon the knowndata sequence position indicator whether the frequency offset estimationvalue being outputted from the multiplier 1308 is valid. If thecontroller 1310 determines that the frequency offset estimation value isvalid, the multiplexer 1309 selects the output of the multiplier 1308.Alternatively, if the controller 1310 determines that the frequencyoffset estimation value is invalid, the controller 1310 generates acontrol signal so that the multiplexer 1309 selects ‘0’. At this point,it is preferable that the input signals stored in the first N symbolbuffer 1301 and in the second N symbol buffer 1303 correspond to signalseach being transmitted by the same known data and passing through almostthe same channel. Otherwise, due to the influence of the transmissionchannel, the frequency offset estimating performance may be largelydeteriorated.

Further, the values N and K of the frequency offset estimator 1092(shown in FIG. 40) may be diversely decided. This is because aparticular portion of the known data that are identically repeated maybe used herein. For example, when the data having the structuredescribed in FIG. 37 are being transmitted, N may be set as BS (i.e.,N=BS), and K may be set as (AS+BS) (i.e., K=AS+BS)). The frequencyoffset estimation value range of the frequency offset estimator 1092 isdecided in accordance with the value K. If the value K is large, thenthe frequency offset estimation value range becomes smaller.Alternatively, if the value K is small, then the frequency offsetestimation value range becomes larger. Therefore, when the data havingthe structure of FIG. 37 is transmitted, and if the repetition cycle(AS+BS) of the known data is long, then the frequency offset estimationvalue range becomes smaller.

In this case, even if the initial frequency offset is estimated by theknown sequence detector and initial frequency offset estimator 1004-1,and if the estimated value is compensated by the second multiplier 1050,the remaining frequency offset after being compensated will exceed theestimation range of the frequency offset estimator 1092. In order toovercome such problems, the known data sequence that is regularlytransmitted may be configured of a repetition of a same data portion byusing a cyclic extension process. For example, if the known datasequence shown in FIG. 37 is configured of two identical portions havingthe length of BS/2, then the N and K values of the frequency offsetestimator 1092 (shown in FIG. 40) may be respectively set as B/2 and B/2(i.e., N=BS/2 and K=BS/2). In this case, the estimation value range maybecome larger than when using repeated known data.

Meanwhile, the known sequence detector and initial frequency offsetestimator 1004-1 detects the place (o position) of the known datasequences that are being periodically or non-periodically transmitted.Simultaneously, the known sequence detector and initial frequency offsetestimator 1004-1 estimates an initial frequency offset during the knownsequence detection process. The known data sequence position indicatordetected by the known sequence detector and initial frequency offsetestimator 1004-1 is outputted to the timing recovery unit 1080, thecarrier recovery unit 1090, and the phase compensator 1110 of thedemodulator 1002, and to the equalizer 1003. Thereafter, the estimatedinitial frequency offset is outputted to the carrier recovery unit 1090.At this point, the known sequence detector and initial frequency offsetestimator 1004-1 may either receive the output of the matched filter1060 or receive the output of the resampler 1040. This may be optionallydecided depending upon the design of the system designer. Herein, thefrequency offset estimator shown in FIG. 40 may be directly applied inthe known sequence detector and initial frequency offset estimator1004-1 or in the phase compensator 1110 of the frequency offsetestimator.

FIG. 41 illustrates a detailed block diagram showing a known sequencedetector and initial frequency offset estimator according to anembodiment of the present invention. More specifically, FIG. 41illustrates an example of an initial frequency offset being estimatedalong with the known sequence position indicator. Herein, FIG. 41 showsan example of an inputted signal being oversampled to N times of itsinitial state. In other words, N represents the sampling rate of areceived signal. Referring to FIG. 41, the known sequence detector andinitial frequency offset estimator includes N number of partialcorrelators 1411 to 141N configured in parallel, a known data placedetector and frequency offset decider 1420, a known data extractor 1430,a buffer 1440, a multiplier 1450, a NCO 1460, a frequency offsetestimator 1470, and an adder 1480. Herein, the first partial correlator1411 consists of a 1/N decimator, and a partial correlator. The secondpartial correlator 1412 consists of a 1 sample delay, a 1/N decimator,and a partial correlator. And, the N^(th) partial correlator 141Nconsists of a N−1 sample delay, a 1/N decimator, and a partialcorrelator. These are used to match (or identify) the phase of each ofthe samples within the oversampled symbol with the phase of the original(or initial) symbol, and to decimate the samples of the remainingphases, thereby performing partial correlation on each sample. Morespecifically, the input signal is decimated at a rate of 1/N for eachsampling phase, so as to pass through each partial correlator.

For example, when the input signal is oversampled to 2 times (i.e., whenN=2), this indicates that two samples are included in one signal. Inthis case, two partial correlators (e.g., 1411 and 1412) are required,and each 1/N decimator becomes a ½ decimator. At this point, the 1/Ndecimator of the first partial correlator 1411 decimates (or removes),among the input samples, the samples located in-between symbol places(or positions). Then, the corresponding 1/N decimator outputs thedecimated sample to the partial correlator. Furthermore, the 1 sampledelay of the second partial correlator 1412 delays the input sample by 1sample (i.e., performs a 1 sample delay on the input sample) and outputsthe delayed input sample to the 1/N decimator. Subsequently, among thesamples inputted from the 1 sample delay, the 1/N decimator of thesecond partial correlator 1412 decimates (or removes) the sampleslocated in-between symbol places (or positions). Thereafter, thecorresponding 1/N decimator outputs the decimated sample to the partialcorrelator.

After each predetermined period of the VSB symbol, each of the partialcorrelators outputs a correlation value and an estimation value of thecoarse frequency offset estimated at that particular moment to the knowndata place detector and frequency offset decider 1420. The known dataplace detector and frequency offset decider 1420 stores the output ofthe partial correlators corresponding to each sampling phase during adata group cycle or a pre-decided cycle. Thereafter, the known dataplace detector and frequency offset decider 1420 decides a position (orplace) corresponding to the highest correlation value, among the storedvalues, as the place (or position) for receiving the known data.Simultaneously, the known data place detector and frequency offsetdecider 1420 finally decides the estimation value of the frequencyoffset estimated at the moment corresponding to the highest correlationvalue as the coarse frequency offset value of the receiving system. Atthis point, the known sequence position indicator is inputted to theknown data extractor 1430, the timing recovery unit 1080, the carrierrecovery unit 1090, the phase compensator 1110, and the equalizer 1003,and the coarse frequency offset is inputted to the adder 1480 and theNCO 1460.

In the meantime, while the N numbers of partial correlators 1411 to 141Ndetect the known data place (or known sequence position) and estimatethe coarse frequency offset, the buffer 1440 temporarily stores thereceived data and outputs the temporarily stored data to the known dataextractor 1430. The known data extractor 1430 uses the known sequenceposition indicator, which is outputted from the known data placedetector and frequency offset decider 1420, so as to extract the knowndata from the output of the buffer 1440. Thereafter, the known dataextractor 1430 outputs the extracted data to the multiplier 1450. TheNCO 1460 generates a complex signal corresponding to the coarsefrequency offset being outputted from the known data place detector andfrequency offset decider 1420. Then, the NCO 1460 outputs the generatedcomplex signal to the multiplier 1450.

The multiplier 1450 multiplies the complex signal of the NCO 1460 to theknown data being outputted from the known data extractor 1430, therebyoutputting the known data having the coarse frequency offset compensatedto the frequency offset estimator 1470. The frequency offset estimator1470 estimates a fine frequency offset from the known data having thecoarse frequency offset compensated. Subsequently, the frequency offsetestimator 1470 outputs the estimated fine frequency offset to the adder1480. The adder 1480 adds the coarse frequency offset to the finefrequency offset. Thereafter, the adder 1480 decides the added result asa final initial frequency offset, which is then outputted to the adder1095 of the carrier recovery unit 1090 included in the demodulator 1002.More specifically, during the process of acquiring initialsynchronization, the present invention may estimate and use the coarsefrequency offset as well as the fine frequency offset, thereby enhancingthe estimation performance of the initial frequency offset.

It is assumed that the known data is inserted within the data group andthen transmitted, as shown in FIG. 5. Then, the known sequence detectorand initial frequency offset estimator 1004-1 may use the known datathat have been additionally inserted between the A1 area and the A2area, so as to estimate the initial frequency offset. The known positionindicator, which was periodically inserted within the A area estimatedby the known sequence detector and initial frequency offset estimator1004-1, is inputted to the timing error detector 1083 of the timingerror recovery unit 1080, to the frequency offset estimator 1092 of thecarrier recovery unit 1090, to the frequency offset estimator 1112 ofthe phase compensator 1110, and to the equalizer 1003.

FIG. 42 illustrates a block diagram showing the structure of one of thepartial correlators shown in FIG. 41. During the step of detecting knowndata, since a frequency offset is included in the received signal, eachpartial correlator divides the known data, which is known according toan agreement between the transmitting system and the receiving system,to K number of parts each having an L symbol length, thereby correlatingeach divided part with the corresponding part of the received signal. Inorder to do so, each partial correlator includes K number of phase andsize detector 1511 to 151K each formed in parallel, an adder 1520, and acoarse frequency offset estimator 1530.

The first phase and size detector 1511 includes an L symbol buffer1511-2, a multiplier 1511-3, an accumulator 1511-4, and a squarer1511-5. Herein, the first phase and size detector 1511 calculates thecorrelation value of the known data having a first L symbol length amongthe K number of sections. Also, the second phase and size detector 1512includes an L symbol delay 1512-1, an L symbol buffer 1512-2, amultiplier 1512-3, an accumulator 1512-4, and a squarer 1512-5. Herein,the second phase and size detector 1512 calculates the correlation valueof the known data having a second L symbol length among the K number ofsections. Finally, the N^(th) phase and size detector 151K includes a(K−1)L symbol delay 151K-1, an L symbol buffer 151K-2, a multiplier151K-3, an accumulator 151K-4, and a squarer 151K-5. Herein, the N^(th)phase and size detector 151K calculates the correlation value of theknown data having an N^(th) L symbol length among the K number ofsections.

Referring to FIG. 42, {P₀, P₁, . . . , P_(KL-1)} each being multipliedwith the received signal in the multiplier represents the known dataknown by both the transmitting system and the receiving system (i.e.,the reference known data generated from the receiving system). And, *represents a complex conjugate. For example, in the first phase and sizedetector 1511, the signal outputted from the 1/N decimator of the firstpartial correlator 1411, shown in FIG. 41, is temporarily stored in theL symbol buffer 1511-2 of the first phase and size detector 1511 andthen inputted to the multiplier 1511-3. The multiplier 1511-3 multipliesthe output of the L symbol buffer 1511-2 with the complex conjugate ofthe known data parts P₀, P₁, . . . , P_(KL-1), each having a first Lsymbol length among the known K number of sections. Then, the multipliedresult is outputted to the accumulator 1511-4. During the L symbolperiod, the accumulator 1511-4 accumulates the output of the multiplier1511-3 and, then, outputs the accumulated value to the squarer 1511-5and the coarse frequency offset estimator 1530. The output of theaccumulator 1511-4 is a correlation value having a phase and a size.Accordingly, the squarer 1511-5 calculates an absolute value of theoutput of the multiplier 1511-4 and squares the calculated absolutevalue, thereby obtaining the size of the correlation value. The obtainedsize is then inputted to the adder 1520.

The adder 1520 adds the output of the squares corresponding to each sizeand phase detector 1511 to 151K. Then, the adder 1520 outputs the addedresult to the known data place detector and frequency offset decider1420. Also, the coarse frequency offset estimator 1530 receives theoutput of the accumulator corresponding to each size and phase detector1511 to 151K, so as to estimate the coarse frequency offset at eachcorresponding sampling phase. Thereafter, the coarse frequency offsetestimator 1530 outputs the estimated offset value to the known dataplace detector and frequency offset decider 1420.

When the K number of inputs that are outputted from the accumulator ofeach phase and size detector 1511 to 151K are each referred to as {Z₀,Z₁, . . . , Z_(K-1)}, the output of the coarse frequency offsetestimator 1530 may be obtained by using Equation 7 shown below.

$\begin{matrix}{\omega_{0} = {\frac{1}{L}\arg \left\{ {\sum\limits_{n = 1}^{K - 1}\; {\left( \frac{Z_{n}}{\left| Z_{n} \right|} \right)\left( \frac{Z_{n - 1}}{\left| Z_{n - 1} \right|} \right)^{*}}} \right\}}} & {{Equation}\mspace{14mu} 7}\end{matrix}$

The known data place detector and frequency offset decider 1420 storesthe output of the partial correlator corresponding to each samplingphase during an enhanced data group cycle or a pre-decided cycle. Then,among the stored correlation values, the known data place detector andfrequency offset decider 1420 decides the place (or position)corresponding to the highest correlation value as the place forreceiving the known data.

Furthermore, the known data place detector and frequency offset decider1420 decides the estimated value of the frequency offset taken (orestimated) at the point of the highest correlation value as the coarsefrequency offset value of the receiving system. For example, if theoutput of the partial correlator corresponding to the second partialcorrelator 1412 is the highest value, the place corresponding to thehighest value is decided as the known data place. Thereafter, the coarsefrequency offset estimated by the second partial correlator 1412 isdecided as the final coarse frequency offset, which is then outputted tothe demodulator 1002.

Meanwhile, the output of the second multiplier 1050 is temporarilystored in the decimator 1081 and the buffer 1082 both included in thetiming recovery unit 1080. Subsequently, the temporarily stored outputdata are inputted to the timing error detector 1083 through thedecimator 1081 and the buffer 1082. Assuming that the output of thesecond multiplier 1050 is oversampled to N times its initial state, thedecimator 1081 decimates the output of the second multiplier 1050 at adecimation rate of 1/N. Then, the 1/N-decimated data are inputted to thebuffer 1082. In other words, the decimator 1081 performs decimation onthe input signal in accordance with a VSB symbol cycle. Furthermore, thedecimator 1081 may also receive the output of the matched filter 1060instead of the output of the second multiplier 1050.

The timing error detector 1083 uses the data prior to or after beingprocessed with matched-filtering and the known sequence positionindicator outputted from the known data detector and initial frequencyoffset estimator 1004-1 in order to detect a timing error. Thereafter,the detected timing error is outputted to the loop filter 1084.Accordingly, the detected timing error information is obtained onceduring each repetition cycle of the known data sequence.

For example, if a known data sequence having the same pattern isperiodically inserted and transmitted, as shown in FIG. 37, the timingerror detector 1083 may use the known data in order to detect the timingerror. There exists a plurality of methods for detecting timing error byusing the known data.

In the example of the present invention, the timing error may bedetected by using a correlation characteristic between the known dataand the received data in the time domain, the known data being alreadyknown in accordance with a pre-arranged agreement between thetransmitting system and the receiving system. The timing error may alsobe detected by using the correlation characteristic of the two knowndata types being received in the frequency domain. Thus, the detectedtiming error is outputted. In another example, a spectral lining methodmay be applied in order to detect the timing error. Herein, the spectrallining method corresponds to a method of detecting timing error by usingsidebands of the spectrum included in the received signal.

The loop filter 1084 filters the timing error detected by the timingerror detector 1083 and, then, outputs the filtered timing error to theholder 1085.

The holder 1085 holds (or maintains) the timing error filtered andoutputted from the loop filter 1084 during a pre-determined known datasequence cycle period and outputs the processed timing error to the NCO1086. Herein, the order of positioning of the loop filter 1084 and theholder 1085 may be switched with one another. In additionally, thefunction of the holder 1085 may be included in the loop filter 1084,and, accordingly, the holder 1085 may be omitted.

The NCO 1086 accumulates the timing error outputted from the holder1085. Thereafter, the NCO 1086 outputs the phase element (i.e., asampling clock) of the accumulated timing error to the resampler 1040,thereby adjusting the sampling timing of the resampler 1040.

FIG. 43 illustrates an example of the timing recovery unit included inthe demodulator 1002 shown in FIG. 36. Referring to FIG. 43, the timingrecovery unit 1080 includes a first timing error detector 1611, a secondtiming error detector 1612, a multiplexer 1613, a loop-filter 1614, andan NCO 1615. The timing recovery unit 1080 would be beneficial when theinput signal is divided into a first area in which known data having apredetermined length are inserted at predetermined position(s) and asecond area that includes no known data. Assuming that the first timingerror detector 1611 detects a first timing error using a sideband of aspectrum of an input signal and the second timing error detector 1612detects a second timing error using the known data, the multiplexer 1613can output the first timing error for the first area and can output thesecond timing error for the second area. The multiplexer 1613 may outputboth of the first and second timing errors for the first area in whichthe known data are inserted. By using the known data a more reliabletiming error can be detected and the performance of the timing recoveryunit 1080 can be enhanced.

This disclosure describes two ways of detecting a timing error. One wayis to detect a timing error using correlation in the time domain betweenknown data pre-known to a transmitting system and a receiving system(reference known data) and the known data actually received by thereceiving system, and the other way is to detect a timing error usingcorrelation in the frequency domain between two known data actuallyreceived by the receiving system. In FIG. 44, a timing error is detectedby calculating correlation between the reference known data pre-known toand generated by the receiving system and the known data actuallyreceived. In FIG. 44, correlation between an entire portion of thereference know data sequence and an entire portion of the received knowndata sequence is calculated. The correlation output has a peak value atthe end of each known data sequence actually received.

In FIG. 45, a timing error is detected by calculating correlation valuesbetween divided portions of the reference known data sequence anddivided portions of the received known data sequence, respectively. Thecorrelation output has a peak value at the end of each divided portionof the received known data sequence. The correlation values may be addedas a total correlation value as shown FIG. 45, and the total correlationvalue can be used to calculate the timing error. When an entire portionof the received known data is used for correlation calculation, thetiming error can be obtained for each data block. If the correlationlevel of the entire portion of the known data sequence is low, a moreprecise correlation can be obtained by using divided portions of theknown data sequence as shown in FIG. 45.

The use of a final correlation value which is obtained based upon aplurality of correlation values of divided portions of a received knowndata sequence may reduce the carrier frequency error. In addition, theprocess time for the timing recovery can be greatly reduced when theplurality of correlation values are used to calculate the timing error.For example, when the reference known data sequence which is pre-knownto the transmitting system and receiving system is divided into Kportions, K correlation values between the K portions of the referenceknown data sequence and the corresponding divided portions of thereceived known data sequence can be calculated, or any combination(s) ofthe correlation values can be used. Therefore, the period of the timingerror detection can be reduced when the divided portions of the knowndata sequence are used instead of the entire portion of the sequence.

The timing error can be calculated from the peak value of thecorrelation values. The timing error is obtained for each data block ifan entire portion of the known data sequence is used as shown in FIG.46. On the other hand, if K divided portions of the known data sequenceare used for correlation calculation, K correlation values andcorresponding peak values can be obtained. This indicates that thetiming error can be detected K times.

A method of detecting a timing error using the correlation between thereference known data and the received known data shown will now bedescribed in more detail. FIG. 46 illustrates correlation values betweenthe reference known data and the received known data. The correlationvalues correspond to data samples sampled at a rate two times greaterthan the symbol clock. When the random data effect is minimized andthere is no timing clock error, the correlation values between thereference known data and the received known data are symmetrical.However, if a timing phase error exists, the correlation values adjacentto the peak value are not symmetrical as shown in FIG. 46. Therefore,the timing error can be obtained by using a difference (timing phaseerror shown in FIG. 46) between the correlation values before and afterthe peak value.

FIG. 47 illustrates an example of the timing error detector shown inFIG. 43. The timing error detector includes a correlator 1701, a downsampler 1702, an absolute value calculator 1703, a delay 1704, and asubtractor 1705. The correlator 1701 receives a known data sequencesampled at a rate at least two times higher than the symbol clockfrequency and calculates the correlation values between the receivedknown data sequence and a reference known data sequence. The downsampler 1702 performs down sampling on the correlation values andobtains samples having a symbol frequency. For example, if the datainputted to the correlator 1701 is pre-sampled at a sampling rate of 2,then the down sampler 1702 performs down sampling at a rate of ½ toobtain samples having the symbol frequency. The absolute valuecalculator 1703 calculates absolute values (or square values) of thedown-sampled correlation values. These absolute values are inputted tothe delay 1704 and the subtractor 1705. The delay 1704 delays theabsolute values for a symbol and the subtractor then outputs a timingerror by subtracting the delayed absolute value from the valuesoutputted from the absolute value calculator 1703.

The arrangement of the correlator 1701, the down sampler 1702, theabsolute value calculator 1703, and the delay 1704, and the subtractor1705 shown in FIG. 47 can be modified. For example, the timing phaseerror can be calculated in the order of the down sampler 1702, thecorrelator 1701, and the absolute value calculator 1703, or in the orderof the correlator 1701, the absolute value calculator 1703, and the downsampler 1702.

The timing error can also be obtained using the frequency characteristicof the known data. When there is a timing frequency error, a phase ofthe input signal increases at a fixed slope as the frequency of thesignal increases and this slope is different for current and next datablock. Therefore, the timing error can be calculated based on thefrequency characteristic of two different known data blocks. In FIG. 48,a current known data sequence (right) and a previous known data sequence(left) are converted into first and second frequency domain signals,respectively, using a Fast Fourier Transform (FFT) algorithm. Theconjugate value of the first frequency domain signal is then multipliedwith the second frequency domain signal in order to obtain thecorrelation value between two frequency domain signals. In other words,the correlation between the frequency value of the previous known datasequence and the frequency value of the current known data sequence isused to detect a phase change between the known data blocks for eachfrequency. In this way the phase distortion of a channel can beeliminated.

The frequency response of a complex VSB signal does not have a fullsymmetric distribution as shown in FIG. 46. Rather, its distribution isa left or right half of the distribution and the frequency domaincorrelation values also have a half distribution. In order to the phasedifference between the frequency domain correlation values, thefrequency domain having the correlation values can be divided into twosub-areas and a phase of a combined correlation value in each sub-areacan be obtained. Thereafter, the difference between the phases ofsub-areas can be used to calculate a timing frequency error. When aphase of a combined correlation values is used for each frequency, themagnitude of each correlation value is proportional to reliability and aphase component of each correlation value is reflected to the finalphase component in proportion to the magnitude.

FIG. 49 illustrates another example of the timing error detector shownin FIG. 43. The timing error detector shown in FIG. 49 includes a FastFourier Transform (FFT) unit 1801, a first delay 1802, a conjugator1803, a multiplier 1804, an accumulator (adder) 1805, a phase detector1806, a second delay 1807, and a subtractor 1808. The first delay 1802delays for one data block and the second delay 1807 delays for ¼ datablock. One data block includes a frequency response of a sequence of Nknown data symbol sequences. When a known data region is known and thedata symbols are received, the FFT unit 1801 converts complex values ofconsecutive N known data symbol sequences into complex values in thefrequency domain. The first delay 1802 delays the frequency domaincomplex values for a time corresponding to one data block, and theconjugator 1803 generate conjugate values of the delayed complex values.The multiplier 1804 multiplies the current block of known data outputtedfrom the FFT unit 1801 with the previous block of known data outputtedfrom the conjugator 1803. The output of the multiplier 1804 representsfrequency region correlation values within a known data block.

Since the complex VSB data exist only on a half of the frequency domain,the accumulator 1805 divides a data region in the known data block intotwo sub-regions, and accumulates correlation values for each sub-region.The phase detector 1806 detects a phase of the accumulated correlationvalue for each sub-region. The second delay 1807 delays the detectedphase for a time corresponding to a ¼ data block. The subtractor 1808obtains a phase difference between the delayed phase and the phaseoutputted from the accumulator 1806 and outputs the phase difference asa timing frequency error.

In the method of calculating a timing error by using a peak ofcorrelation between the reference known data and the received known datain the time domain, the contribution of the correlation values mayaffect a channel when the channel is a multi path channel. However, thiscan be greatly eliminated if the timing error is obtained using thecorrelation between two received known data. In addition, the timingerror can be detected using an entire portion of the known data sequenceinserted by the transmitting system, or it can be detected using aportion of the known data sequence which is robust to random or noisedata.

Meanwhile, the DC remover 1070 removes pilot tone signal (i.e., DCsignal), which has been inserted by the transmitting system, from thematched-filtered signal. Thereafter, the DC remover 1070 outputs theprocessed signal to the phase compensator 1110.

FIG. 50 illustrates a detailed block diagram of a DC remover accordingto an embodiment of the present invention. Herein, identical signalprocessing processes are performed on each of a real number element (orin-phase (I)) and an imaginary number element (or a quadrature (Q)) ofthe inputted complex signal, thereby estimating and removing the DCvalue of each element. In order to do so, the DC remover shown in FIG.50 includes a first DC estimator and remover 1900 and a second DCestimator and remover 1950. Herein, the first DC estimator and remover1900 includes an R sample buffer 1901, a DC estimator 1902, an M sampleholder 1903, a C sample delay 1904, and a subtractor 1905. Herein, thefirst DC estimator and remover 1900 estimates and removes the DC of thereal number element (i.e., an in-phase DC). Furthermore, the second DCestimator and remover 1950 includes an R sample buffer 1951, a DCestimator 1952, an M sample holder 1953, a C sample delay 1954, and asubtractor 1955. The second DC estimator and remover 1950 estimates andremoves the DC of the imaginary number element (i.e., a quadrature DC).In the present invention, the first DC estimator and remover 1900 andthe second DC estimator and remover 1950 may receive different inputsignals. However, each DC estimator and remover 1900 and 1950 has thesame structure. Therefore, a detailed description of the first DCestimator and remover 1900 will be presented herein, and the second DCestimator and remover 1950 will be omitted for simplicity.

More specifically, the in-phase signal matched-filtered by the matchedfilter 1060 is inputted to the R sample buffer 1901 of the first DCestimator and remover 1900 within the DC remover 1070 and is thenstored. The R sample buffer 1901 is a buffer having the length of Rsample. Herein, the output of the R sample buffer 1901 is inputted tothe DC estimator 1902 and the C sample delay 1904. The DC estimator 1902uses the data having the length of R sample, which are outputted fromthe buffer 1901, so as to estimate the DC value by using Equation 8shown below.

$\begin{matrix}{{y\lbrack n\rbrack} = {\frac{1}{R}{\sum\limits_{k = 0}^{R - 1}\; {x\left\lbrack {k + {M*n}} \right\rbrack}}}} & {{Equation}\mspace{14mu} 8}\end{matrix}$

In the above-described Equation 8, x[n] represents the inputted sampledata stored in the buffer 1901. And, y[n] indicates the DC estimationvalue. More specifically, the DC estimator 1902 accumulates R number ofsample data stored in the buffer 1901 and estimates the DC value bydividing the accumulated value by R. At this point, the stored inputsample data set is shifted as much as M sample. Herein, the DCestimation value is outputted once every M samples.

FIG. 51 illustrates a shifting of the input sample data used for DCestimation. For example, when M is equal to 1 (i.e., M=1), the DCestimator 1902 estimates the DC value each time a sample is shifted tothe buffer 1901. Accordingly, each estimated result is outputted foreach sample. If M is equal to R (i.e., M=R), the DC estimator 1902estimates the DC value each time R number of samples are shifted to thebuffer 1901. Accordingly, each estimated result is outputted for eachcycle of R samples. Therefore, in this case, the DC estimator 1902corresponds to a DC estimator that operates in a block unit of Rsamples. Herein, any value within the range of 1 and R may correspond tothe value M.

As described above, since the output of the DC estimator 1902 isoutputted after each cycle of M samples, the M sample holder 1903 holdsthe DC value estimated from the DC estimator 1902 for a period of Msamples. Then, the estimated DC value is outputted to the subtractor1905. Also, the C sample delay 1904 delays the input sample data storedin the buffer 1901 by C samples, which are then outputted to thesubtractor 1905. The subtractor 1905 subtracts the output of the Msample holder 1903 from the output of the C sample delay 1904.Thereafter, the subtractor 1905 outputs the signal having the in-phaseDC removed.

Herein, the C sample delay 1904 decides which portion of the inputsample data is to be compensated with the output of the DC estimator1902. More specifically, the DC estimator and remover 1900 may bedivided into a DC estimator 1902 for estimating the DC and thesubtractor for compensating the input sample data within the estimatedDC value. At this point, the C sample delay 1904 decides which portionof the input sample data is to be compensated with the estimated DCvalue. For example, when C is equal to 0 (i.e., C=0), the beginning ofthe R samples is compensated with the estimated DC value obtained byusing R samples. Alternatively, when C is equal to R (i.e., C=R), theend of the R samples is compensated with the estimated DC value obtainedby using R samples. Similarly, the data having the DC removed areinputted to the buffer 1111 and the frequency offset estimator 1112 ofthe phase compensator 1110.

Meanwhile, FIG. 52 illustrates a detailed block diagram of a DC removeraccording to another embodiment of the present invention. Herein,identical signal processing processes are performed on each of a realnumber element (or in-phase (I)) and an imaginary number element (or aquadrature (Q)) of the inputted complex signal, thereby estimating andremoving the DC value of each element. In order to do so, the DC removershown in FIG. 52 includes a first DC estimator and remover 2100 and asecond DC estimator and remover 2150. FIG. 52 corresponds to an infiniteimpulse response (IIR) structure.

Herein, the first DC estimator and remover 2100 includes a multiplier2101, an adder 2102, an 1 sample delay 2103, a multiplier 2104, a Csample delay 2105, and a subtractor 2106. Also, the second DC estimatorand remover 2150 includes a multiplier 2151, an adder 2152, an 1 sampledelay 2153, a multiplier 2154, a C sample delay 2155, and a subtractor2156. In the present invention, the first DC estimator and remover 2100and the second DC estimator and remover 2150 may receive different inputsignals. However, each DC estimator and remover 2100 and 2150 has thesame structure. Therefore, a detailed description of the first DCestimator and remover 2100 will be presented herein, and the second DCestimator and remover 2150 will be omitted for simplicity.

More specifically, the in-phase signal matched-filtered by the matchedfilter 1060 is inputted to the multiplier 2101 and the C sample delay2105 of the first DC estimator and remover 2100 within the DC remover1070. The multiplier 2101 multiplies a pre-determined constant α to thein-phase signal that is being inputted. Then, the multiplier 2101outputs the multiplied result to the adder 2102. The adder 2102 adds theoutput of the multiplier 2101 to the output of the multiplier 2104 thatis being fed-back. Thereafter, the adder 2102 outputs the added resultto the 1 sample delay 2103 and the subtractor 2106. More specifically,the output of the adder 2102 corresponds to the estimated in-phase DCvalue.

The 1 sample delay 2103 delays the estimated DC value by 1 sample andoutputs the DC value delayed by 1 sample to the multiplier 2104. Themultiplier 2104 multiplies a pre-determined constant (1−α) to the DCvalue delayed by 1 sample. Then, the multiplier 2104 feeds-back themultiplied result to the adder 2102.

Subsequently, the C sample delay 2105 delays the in-phase sample data byC samples and, then, outputs the delayed in-phase sample data to thesubtractor 2106. The subtractor 2106 subtracts the output of the adder2102 from the output of the C sample delay 2105, thereby outputting thesignal having the in-phase DC removed therefrom.

Similarly, the data having the DC removed are inputted to the buffer1111 and the frequency offset estimator 1112 of the phase compensator1110 of FIG. 39.

The frequency offset estimator 1112 uses the known sequence positionindicator outputted from the known sequence detector and initialfrequency offset estimator 1004-1 in order to estimate the frequencyoffset from the known data sequence that is being inputted, the knowndata sequence having the DC removed by the DC remover 1070. Then, thefrequency offset estimator 1112 outputs the estimated frequency offsetto the holder 1113. Similarly, the frequency offset estimation value isobtained at each repetition cycle of the known data sequence.

Therefore, the holder 1113 holds the frequency offset estimation valueduring a cycle period of the known data sequence and then outputs thefrequency offset estimation value to the NCO 1114. The NCO 1114generates a complex signal corresponding to the frequency offset held bythe holder 1113 and outputs the generated complex signal to themultiplier 1115.

The multiplier 1115 multiplies the complex signal outputted from the NCO1114 to the data being delayed by a set period of time in the buffer1111, thereby compensating the phase change included in the delayeddata. The data having the phase change compensated by the multiplier1115 pass through the decimator 1200 so as to be inputted to theequalizer 1003. At this point, since the frequency offset estimated bythe frequency offset estimator 1112 of the phase compensator 1110 doesnot pass through the loop filter, the estimated frequency offsetindicates the phase difference between the known data sequences. Inother words, the estimated frequency offset indicates a phase offset.

Channel Equalizer

The demodulated data using the known data in the demodulator 1002 isinputted to the channel equalizer 1003. The demodulated data is inputtedto the known sequence detector 1004.

The equalizer 1003 may perform channel equalization by using a pluralityof methods. An example of estimating a channel impulse response (CIR) soas to perform channel equalization will be given in the description ofthe present invention. Most particularly, an example of estimating theCIR in accordance with each region within the data group, which ishierarchically divided and transmitted from the transmitting system, andapplying each CIR differently will also be described herein.Furthermore, by using the known data, the place and contents of which isknown in accordance with an agreement between the transmitting systemand the receiving system, and/or the field synchronization data, so asto estimate the CIR, the present invention may be able to performchannel equalization with more stability.

Herein, the data group that is inputted for the equalization process isdivided into regions A to D, as shown in FIG. 5. More specifically, inthe example of the present invention, each region A, B, C, and D arefurther divided into MPH blocks B4 to B7, MPH blocks B3 and B8, MPHblocks B2 and B9, MPH blocks B1 and B10, respectively.

More specifically, a data group can be assigned and transmitted amaximum the number of 4 in a VSB frame in the transmitting system. Inthis case, all data group do not include field synchronization data. Inthe present invention, the data group including the fieldsynchronization data performs channel-equalization using the fieldsynchronization data and known data. And the data group not includingthe field synchronization data performs channel-equalization using theknown data. For example, the data of the MPH block B3 including thefield synchronization data performs channel-equalization using the CIRcalculated from the field synchronization data area and the CIRcalculated from the first known data area. Also, the data of the MPHblocks B1 and B2 performs channel-equalization using the CIR calculatedfrom the field synchronization data area and the CIR calculated from thefirst known data area. Meanwhile, the data of the MPH blocks B4 to B6not including the field synchronization data performschannel-equalization using CIRS calculated from the first known dataarea and the third known data area.

As described above, the present invention uses the CIR estimated fromthe field synchronization data and the known data sequences in order toperform channel equalization on data within the data group. At thispoint, each of the estimated CIRs may be directly used in accordancewith the characteristics of each region within the data group.Alternatively, a plurality of the estimated CIRs may also be eitherinterpolated or extrapolated so as to create a new CIR, which is thenused for the channel equalization process.

Herein, when a value F(Q) of a function F(x) at a particular point Q anda value F(S) of the function F(x) at another particular point S areknown, interpolation refers to estimating a function value of a pointwithin the section between points Q and S. Linear interpolationcorresponds to the simplest form among a wide range of interpolationoperations. The linear interpolation described herein is merelyexemplary among a wide range of possible interpolation methods. And,therefore, the present invention is not limited only to the examples setforth herein.

Alternatively, when a value F(Q) of a function F(x) at a particularpoint Q and a value F(S) of the function F(x) at another particularpoint S are known, extrapolation refers to estimating a function valueof a point outside of the section between points Q and S. Linearextrapolation is the simplest form among a wide range of extrapolationoperations. Similarly, the linear extrapolation described herein ismerely exemplary among a wide range of possible extrapolation methods.And, therefore, the present invention is not limited only to theexamples set forth herein.

FIG. 53 illustrates a block diagram of a channel equalizer according toanother embodiment of the present invention. Herein, by estimating andcompensating a remaining carrier phase error from a channel-equalizedsignal, the receiving system of the present invention may be enhanced.Referring to FIG. 53, the channel equalizer includes a first frequencydomain converter 3100, a channel estimator 3110, a second frequencydomain converter 3121, a coefficient calculator 3122, a distortioncompensator 3130, a time domain converter 3140, a remaining carrierphase error remover 3150, a noise canceller (NC) 3160, and a decisionunit 3170.

Herein, the first frequency domain converter 3100 includes an overlapunit 3101 overlapping inputted data, and a fast fourier transform (FFT)unit 3102 converting the data outputted from the overlap unit 3101 tofrequency domain data.

The channel estimator 3110 includes a CIR estimator, a phase compensator3112, a pre-CIR cleaner 3113, CIR interpolator/extrapolator 3114, apost-CIR cleaner, and a zero-padding unit.

The second frequency domain converter 3121 includes a fast fouriertransform (FFT) unit converting the CIR being outputted from the channelestimator 3110 to frequency domain CIR.

The time domain converter 3140 includes an IFFT unit 3141 converting thedata having the distortion compensated by the distortion compensator3130 to time domain data, and a save unit 3142 extracting only validdata from the data outputted from the IFFT unit 3141.

The remaining carrier phase error remover 3150 includes an errorcompensator 3151 removing the remaining carrier phase error included inthe channel equalized data, and a remaining carrier phase errorestimator 3152 using the channel equalized data and the decision data ofthe decision unit 3170 so as to estimate the remaining carrier phaseerror, thereby outputting the estimated error to the error compensator3151. Herein, any device performing complex number multiplication may beused as the distortion compensator 3130 and the error compensator 3151.

At this point, since the received data correspond to data modulated toVSB type data, 8-level scattered data exist only in the real numberelement. Therefore, referring to FIG. 53, all of the signals used in thenoise canceller 3160 and the decision unit 3170 correspond to realnumber (or in-phase) signals. However, in order to estimate andcompensate the remaining carrier phase error and the phase noise, bothreal number (in-phase) element and imaginary number (quadrature) elementare required. Therefore, the remaining carrier phase error remover 3150receives and uses the quadrature element as well as the in-phaseelement. Generally, prior to performing the channel equalizationprocess, the demodulator 902 within the receiving system performsfrequency and phase recovery of the carrier. However, if a remainingcarrier phase error that is not sufficiently compensated is inputted tothe channel equalizer, the performance of the channel equalizer may bedeteriorated. Particularly, in a dynamic channel environment, theremaining carrier phase error may be larger than in a static channelenvironment due to the frequent and sudden channel changes. Eventually,this acts as an important factor that deteriorates the receivingperformance of the present invention.

Furthermore, a local oscillator (not shown) included in the receivingsystem should preferably include a single frequency element. However,the local oscillator actually includes the desired frequency elements aswell as other frequency elements. Such unwanted (or undesired) frequencyelements are referred to as phase noise of the local oscillator. Suchphase noise also deteriorates the receiving performance of the presentinvention. It is difficult to compensate such remaining carrier phaseerror and phase noise from the general channel equalizer. Therefore, thepresent invention may enhance the channel equaling performance byincluding a carrier recovery loop (i.e., a remaining carrier phase errorremover 3150) in the channel equalizer, as shown in FIG. 53, in order toremove the remaining carrier phase error and the phase noise.

More specifically, the receiving data demodulated in FIG. 53 areoverlapped by the overlap unit 3101 of the first frequency domainconverter 3100 at a pre-determined overlapping ratio, which are thenoutputted to the FFT unit 3102. The FFT unit 3102 converts theoverlapped time domain data to overlapped frequency domain data throughby processing the data with FFT. Then, the converted data are outputtedto the distortion compensator 3130.

The distortion compensator 3130 performs a complex number multiplicationon the overlapped frequency domain data outputted from the FFT unit 3102included in the first frequency domain converter 3100 and theequalization coefficient calculated from the coefficient calculator3122, thereby compensating the channel distortion of the overlapped dataoutputted from the FFT unit 3102. Thereafter, the compensated data areoutputted to the IFFT unit 3141 of the time domain converter 3140. TheIFFT unit 3141 performs IFFT on the overlapped data having the channeldistortion compensated, thereby converting the overlapped data to timedomain data, which are then outputted to the error compensator 3151 ofthe remaining carrier phase error remover 3150.

The error compensator 3151 multiplies a signal compensating theestimated remaining carrier phase error and phase noise with the validdata extracted from the time domain. Thus, the error compensator 3151removes the remaining carrier phase error and phase noise included inthe valid data.

The data having the remaining carrier phase error compensated by theerror compensator 3151 are outputted to the remaining carrier phaseerror estimator 3152 in order to estimate the remaining carrier phaseerror and phase noise and, at the same time, outputted to the noisecanceller 3160 in order to remove (or cancel) the noise.

The remaining carrier phase error estimator 3152 uses the output data ofthe error compensator 3151 and the decision data of the decision unit3170 to estimate the remaining carrier phase error and phase noise.Thereafter, the remaining carrier phase error estimator 3152 outputs asignal for compensating the estimated remaining carrier phase error andphase noise to the error compensator 3151. In this embodiment of thepresent invention, an inverse number of the estimated remaining carrierphase error and phase noise is outputted as the signal for compensatingthe remaining carrier phase error and phase noise.

FIG. 54 illustrates a detailed block diagram of the remaining carrierphase error estimator 3152 according to an embodiment of the presentinvention. Herein, the remaining carrier phase error estimator 3152includes a phase error detector 3211, a loop filter 3212, a numericallycontrolled oscillator (NCO) 3213, and a conjugator 3214. Referring toFIG. 54, the decision data, the output of the phase error detector 3211,and the output of the loop filter 3212 are all real number signals. And,the output of the error compensator 3151, the output of the NCO 3213,and the output of the conjugator 3214 are all complex number signals.

The phase error detector 3211 receives the output data of the errorcompensator 3151 and the decision data of the decision unit 3170 inorder to estimate the remaining carrier phase error and phase noise.Then, the phase error detector 3211 outputs the estimated remainingcarrier phase error and phase noise to the loop filter.

The loop filter 3212 then filters the remaining carrier phase error andphase noise, thereby outputting the filtered result to the NCO 3213. TheNCO 3213 generates a cosine corresponding to the filtered remainingcarrier phase error and phase noise, which is then outputted to theconjugator 3214.

The conjugator 3214 calculates the conjugate value of the cosine wavegenerated by the NCO 3213. Thereafter, the calculated conjugate value isoutputted to the error compensator 3151. At this point, the output dataof the conjugator 3214 becomes the inverse number of the signalcompensating the remaining carrier phase error and phase noise. In otherwords, the output data of the conjugator 3214 becomes the inverse numberof the remaining carrier phase error and phase noise.

The error compensator 3151 performs complex number multiplication on theequalized data outputted from the time domain converter 3140 and thesignal outputted from the conjugator 3214 and compensating the remainingcarrier phase error and phase noise, thereby removing the remainingcarrier phase error and phase noise included in the equalized data.Meanwhile, the phase error detector 3211 may estimate the remainingcarrier phase error and phase noise by using diverse methods andstructures. According to this embodiment of the present invention, theremaining carrier phase error and phase noise are estimated by using adecision-directed method.

If the remaining carrier phase error and phase noise are not included inthe channel-equalized data, the decision-directed phase error detectoraccording to the present invention uses the fact that only real numbervalues exist in the correlation values between the channel-equalizeddata and the decision data. More specifically, if the remaining carrierphase error and phase noise are not included, and when the input data ofthe phase error detector 3211 are referred to as x_(i)+jx_(q), thecorrelation value between the input data of the phase error detector3211 and the decision data may be obtained by using Equation 9 shownbelow:

E{(x _(i) +jx _(q))({circumflex over (x)} _(i) +j{circumflex over (x)}_(q))*}  Equation 9

At this point, there is no correlation between x_(i) and x_(q).Therefore, the correlation value between x_(i) and x_(q) is equal to 0.Accordingly, if the remaining carrier phase error and phase noise arenot included, only the real number values exist herein. However, if theremaining carrier phase error and phase noise are included, the realnumber element is shown in the imaginary number value, and the imaginarynumber element is shown in the real number value. Thus, in this case,the imaginary number element is shown in the correlation value.Therefore, it can be assumed that the imaginary number portion of thecorrelation value is in proportion with the remaining carrier phaseerror and phase noise. Accordingly, as shown in Equation 10 below, theimaginary number of the correlation value may be used as the remainingcarrier phase error and phase noise.

Phase Error=imag{(x _(i) +jx _(q))({circumflex over (x)} _(i)+j{circumflex over (x)} _(q))*}

Phase Error=x _(q) {circumflex over (x)} _(i) −x _(i) {circumflex over(x)} _(q)  Equation 10

FIG. 55 illustrates a block diagram of a phase error detector 3211obtaining the remaining carrier phase error and phase noise. Herein, thephase error detector 3211 includes a Hilbert converter 3311, a complexnumber configurator 3312, a conjugator 3313, a multiplier 3314, and aphase error output 3315. More specifically, the Hilbert converter 3311creates an imaginary number decision data {circumflex over (x)}_(q) byperforming a Hilbert conversion on the decision value {circumflex over(x)}_(i) of the decision unit 3170. The generated imaginary numberdecision value is then outputted to the complex number configurator3312. The complex number configurator 3312 uses the decision data{circumflex over (x)}_(i) and {circumflex over (x)}_(q) to configure thecomplex number decision data {circumflex over (x)}_(i)+j{circumflex over(x)}_(q), which are then outputted to the conjugator 3313. Theconjugator 3313 conjugates the output of the complex number configurator3312, thereby outputting the conjugated value to the multiplier 3314.The multiplier 3314 performs a complex number multiplication on theoutput data of the error compensator 3151 and the output data{circumflex over (x)}_(i)−j{circumflex over (x)}_(q) of the conjugator3313, thereby obtaining the correlation between the output datax_(i)+jx_(q) of the error compensator 3151 and the decision value{circumflex over (x)}_(i)−j{circumflex over (x)}_(q) of the decisionunit 3170. The correlation data obtained from the multiplier 3314 arethen inputted to the phase error output 3315. The phase error output3315 outputs the imaginary number portion x_(q){circumflex over(x)}_(i)−x_(i){circumflex over (x)}_(q) of the correlation dataoutputted from the multiplier 3314 as the remaining carrier phase errorand phase noise.

The phase error detector shown in FIG. 55 is an example of a pluralityof phase error detecting methods. Therefore, other types of phase errordetectors may be used in the present invention. Therefore, the presentinvention is not limited only to the examples and embodiments presentedin the description of the present invention. Furthermore, according toanother embodiment of the present invention, at least 2 phase errordetectors are combined so as to detect the remaining carrier phase errorand phase noise.

Accordingly, the output of the remaining carrier phase error remover3150 having the detected remaining carrier phase error and phase noiseremoved as described above, is configured of an addition of the original(or initial) signal having the channel equalization, the remainingcarrier phase error and phase noise, and the signal corresponding to awhite noise being amplified to a colored noise during the channelequalization.

Therefore, the noise canceller 3160 receives the output data of theremaining carrier phase error remover 3150 and the decision data of thedecision unit 3170, thereby estimating the colored noise. Then, thenoise canceller 3160 subtracts the estimated colored noise from the datahaving the remaining carrier phase error and phase noise removedtherefrom, thereby removing the noise amplified during the equalizationprocess.

In order to do so, the noise canceller 3160 includes a subtractor and anoise predictor. More specifically, the subtractor subtracts the noisepredicted by the noise predictor from the output data of the residualcarrier phase error estimator 3150. Then, the subtractor outputs thesignal from which amplified noise is cancelled (or removed) for datarecovery and, simultaneously, outputs the same signal to the decisionunit 3170. The noise predictor calculates a noise element by subtractingthe output of the decision unit 3170 from the signal having residualcarrier phase error removed therefrom by the residual carrier phaseerror estimator 3150. Thereafter, the noise predictor uses thecalculated noise element as input data of a filter included in the noisepredictor. Also, the noise predictor uses the filter (not shown) inorder to predict any color noise element included in the output symbolof the residual carrier phase error estimator 3150. Accordingly, thenoise predictor outputs the predicted color noise element to thesubtractor.

The data having the noise removed (or cancelled) by the noise canceller3160 are outputted for the data decoding process and, at the same time,outputted to the decision unit 3170.

The decision unit 3170 selects one of a plurality of pre-determineddecision data sets (e.g., 8 decision data sets) that is most approximateto the output data of the noise canceller 3160, thereby outputting theselected data to the remaining carrier phase error estimator 3152 andthe noise canceller 3160.

Meanwhile, the received data are inputted to the overlap unit 3101 ofthe first frequency domain converter 3100 included in the channelequalizer and, at the same time, inputted to the CIR estimator 3111 ofthe channel estimator 3110.

The CIR estimator 3111 uses a training sequence, for example, data beinginputted during the known data section and the known data in order toestimate the CIR, thereby outputting the estimated CIR to the phasecompensator 3112. If the data to be channel-equalizing is the datawithin the data group including field synchronization data, the trainingsequence using in the CIR estimator 3111 may become the fieldsynchronization data and known data. Meanwhile, if the data to bechannel-equalizing is the data within the data group not including fieldsynchronization data, the training sequence using in the CIR estimator3111 may become only the known data.

For example, the CIR estimator 3111 estimates CIR using the known datacorrespond to reference known data generated during the known datasection by the receiving system in accordance with an agreement betweenthe receiving system and the transmitting system. For this, the CIRestimator 3111 is provided known data position information from theknown sequence detector 1004. Also the CIR estimator 3111 may beprovided field synchronization position information from the knownsequence detector 1004.

Furthermore, in this embodiment of the present invention, the CIRestimator 3111 estimates the CIR by using the least square (LS) method.

The LS estimation method calculates a cross correlation value p betweenthe known data that have passed through the channel during the knowndata section and the known data that are already known by the receivingend. Then, a cross correlation matrix R of the known data is calculated.Subsequently, a matrix operation is performed on R⁻¹·p so that the crosscorrelation portion within the cross correlation value p between thereceived data and the initial known data, thereby estimating the CIR ofthe transmission channel.

The phase compensator 3112 compensates the phase change of the estimatedCIR. Then, the phase compensator 3112 outputs the compensated CIR to thelinear interpolator 3113. At this point, the phase compensator 3112 maycompensate the phase change of the estimated CIR by using a maximumlikelihood method.

More specifically, the remaining carrier phase error and phase noisethat are included in the demodulated received data and, therefore, beinginputted change the phase of the CIR estimated by the CIR estimator 3111at a cycle period of one known data sequence. At this point, if thephase change of the inputted CIR, which is to be used for the linearinterpolation process, is not performed in a linear form due to a highrate of the phase change, the channel equalizing performance of thepresent invention may be deteriorated when the channel is compensated bycalculating the equalization coefficient from the CIR, which isestimated by a linear interpolation method.

Therefore, the present invention removes (or cancels) the amount ofphase change of the CIR estimated by the CIR estimator 3111 so that thedistortion compensator 3130 allows the remaining carrier phase error andphase noise to bypass the distortion compensator 3130 without beingcompensated. Accordingly, the remaining carrier phase error and phasenoise are compensated by the remaining carrier phase error remover 3150.

For this, the present invention removes (or cancels) the amount of phasechange of the CIR estimated by the phase compensator 3112 by using amaximum likelihood method.

The basic idea of the maximum likelihood method relates to estimating aphase element mutually (or commonly) existing in all CIR elements, thento multiply the estimated CIR with an inverse number of the mutual (orcommon) phase element, so that the channel equalizer, and mostparticularly, the distortion compensator 3130 does not compensate themutual phase element.

More specifically, when the mutual phase element is referred to as θ,the phase of the newly estimated CIR is rotated by θ as compared to thepreviously estimated CIR. When the CIR of a point t is referred to ash_(i)(t), the maximum likelihood phase compensation method obtains aphase θ_(ML) corresponding to when h_(i)(t) is rotated by θ, the squaredvalue of the difference between the CIR of h_(i)(t) and the CIR ofh_(i)(t+1), i.e., the CIR of a point (t+1), becomes a minimum value.Herein, when i represents a tap of the estimated CIR, and when Nrepresents a number of taps of the CIR being estimated by the CIRestimator 3111, the value of θ_(ML) is equal to or greater than 0 andequal to or less than N−1. This value may be calculated by usingEquation 11 shown below:

$\begin{matrix}{\theta_{ML} = {\begin{matrix}\min \\\theta\end{matrix}{\sum\limits_{i = 0}^{N - 1}\; {{{{h_{i}(t)}^{j\theta}} - {h_{i}\left( {t + 1} \right)}}}^{2}}}} & {{Equation}\mspace{14mu} 11}\end{matrix}$

Herein, in light of the maximum likelihood method, the mutual phaseelement θ_(ML) is equal to the value of θ, when the right side ofEquation 11 being differentiated with respect to θ is equal to 0. Theabove-described condition is shown in Equation 12 below:

$\begin{matrix}{{\frac{\;}{\theta}{\sum\limits_{i = 0}^{N - 1}\; {{{{h_{i}(t)}^{j\theta}} - {h_{i}\left( {t + 1} \right)}}}^{2}}} = {{\frac{\;}{\theta}{\sum\limits_{i = 0}^{N - 1}{\left( {{{h_{i}(t)}^{j\theta}} - {h_{i}\left( {t + 1} \right)}} \right)\left( {{{h_{i\;}(t)}^{j\theta}} - {h_{i}\left( {t + 1} \right)}} \right)^{*}}}} = {{\frac{\;}{\theta}{\sum\limits_{i = 0}^{N - 1}\left\{ {{{h_{i}(t)}}^{2} + {{h_{i + 1}(t)}}^{2} - {{h_{i}(t)}{h_{i}^{*}\left( {t + 1} \right)}^{j\theta}} - {{h_{i}^{*}(t)}{h_{i}\left( {t + 1} \right)}^{- {j\theta}}}} \right\}}} = {{\sum\limits_{i = 0}^{N - 1}\left\{ {{j\; {h_{i}^{*}(t)}{h_{i}\left( {t + 1} \right)}^{- {j\theta}}} - {j\; {h_{i}^{*}(t)}{h_{i}\left( {t + 1} \right)}^{j\theta}}} \right\}} = {{j{\sum\limits_{i = 0}^{N - 1}{2{Im}\left\{ {{h_{i}^{*}(t)}{h_{i}\left( {t + 1} \right)}^{- {j\theta}}} \right\}}}} = 0}}}}} & {{Equation}\mspace{14mu} 12}\end{matrix}$

The above Equation 12 may be simplified as shown in Equation 13 below:

$\begin{matrix}{{{{Im}\left\{ {^{- {j\theta}}{\sum\limits_{i = 0}^{N - 1}\left\{ {{h_{i}^{*}(t)}{h_{i}\left( {t + 1} \right)}} \right\}}} \right\}} = 0}{\theta_{ML} = {\arg\left( {\sum\limits_{i = 0}^{N - 1}{{h_{i}^{*}(t)}{h_{i}\left( {t + 1} \right)}}} \right)}}} & {{Equation}\mspace{14mu} 13}\end{matrix}$

More specifically, Equation 13 corresponds to the θ_(ML) value that isto be estimated by the argument of the correlation value betweenh_(i)(t) and h_(i)(t+1).

FIG. 56 illustrates a phase compensator according to an embodiment ofthe present invention, wherein the mutual phase element θ_(ML) iscalculated as described above, and wherein the estimated phase elementis compensated at the estimated CIR. Referring to FIG. 56, the phasecompensator includes a correlation calculator 3410, a phase changeestimator 3420, a compensation signal generator 3430, and a multiplier3440.

The correlation calculator 3410 includes a first N symbol buffer 3411,an N symbol delay 3412, a second N symbol buffer 3413, a conjugator3414, and a multiplier 3415. More specifically, the first N symbolbuffer 3411 included in the correlation calculator 3410 is capable ofstoring the data being inputted from the CIR estimator 3111 in symbolunits to a maximum limit of N number of symbols. The symbol data beingtemporarily stored in the first N symbol buffer 3411 are then inputtedto the multiplier 3415 included in the correlation calculator 3410 andto the multiplier 3440.

At the same time, the symbol data being outputted from the CIR estimator3111 are delayed by N symbols from the N symbol delay 3412. Then, thedelayed symbol data pass through the second N symbol buffer 3413 andinputted to the conjugator 3414, so as to be conjugated and theninputted to the multiplier 3415.

The multiplier 3415 multiplies the output of the first N symbol buffer3411 and the output of the conjugator 3414. Then, the multiplier 3415outputs the multiplied result to an accumulator 3421 included in thephase change estimator 3420.

More specifically, the correlation calculator 3410 calculates acorrelation between a current CIR h_(i)(t+1) having the length of N anda previous CIR h_(i)(t) also having the length of N. then, thecorrelation calculator 3410 outputs the calculated correlation value tothe accumulator 3421 of the phase change estimator 3420.

The accumulator 3421 accumulates the correlation values outputted fromthe multiplier 3415 during an N symbol period. Then, the accumulator3421 outputs the accumulated value to the phase detector 3422. The phasedetector 3422 then calculates a mutual phase element θ_(ML) from theoutput of the accumulator 3421 as shown in the above-described Equation11. Thereafter, the calculated θ_(ML) value is outputted to thecompensation signal generator 3430.

The compensation signal generator 3430 outputs a complex signal e^(−jθ)^(ML) having a phase opposite to that of the detected phase as the phasecompensation signal to the multiplier 3440. The multiplier 3440multiplies the current CIR h_(i)(t+1) being outputted from the first Nsymbol buffer 3411 with the phase compensation signal e^(−jθ) ^(ML) ,thereby removing the amount of phase change of the estimated CIR.

The CIR having its phase change compensated, as described above, passesthrough a first cleaner (or pre-CIR cleaner) 3113 or bypasses the firstcleaner 3113, thereby being inputted to a CIR calculator (or CIRinterpolator-extrapolator) 3114. The CIR interpolator-extrapolator 3114either interpolates or extrapolates an estimated CIR, which is thenoutputted to a second cleaner (or post-CIR cleaner) 3115. Herein, theestimated CIR corresponds to a CIR having its phase change compensated.The first cleaner 3113 may or may not operate depending upon whether theCIR interpolator-extrapolator 3114 interpolates or extrapolates theestimated CIR. For example, if the CIR interpolator-extrapolator 3114interpolates the estimated CIR, the first cleaner 3113 does not operate.Conversely, if the CIR interpolator-extrapolator 3114 extrapolates theestimated CIR, the first cleaner 3113 operates.

More specifically, the CIR estimated from the known data includes achannel element that is to be obtained as well as a jitter elementcaused by noise. Since such jitter element deteriorates the performanceof the equalizer, it preferable that a coefficient calculator 3122removes the jitter element before using the estimated CIR. Therefore,according to the embodiment of the present invention, each of the firstand second cleaners 3113 and 3115 removes a portion of the estimated CIRhaving a power level lower than the predetermined threshold value (i.e.,so that the estimated CIR becomes equal to ‘0’). Herein, this removalprocess will be referred to as a “CIR cleaning” process.

The CIR interpolator-extrapolator 3114 performs CIR interpolation bymultiplying a CIR estimated from the CIR estimator 3112 by a coefficientand by multiplying a CIR having its phase change compensated from thephase compensator (or maximum likelihood phase compensator) 3112 byanother coefficient, thereby adding the multiplied values. At thispoint, some of the noise elements of the CIR may be added to oneanother, thereby being cancelled. Therefore, when the CIRinterpolator-extrapolator 3114 performs CIR interpolation, the original(or initial) CIR having noise elements remaining therein. In otherwords, when the CIR interpolator-extrapolator 3114 performs CIRinterpolation, an estimated CIR having its phase change compensated bythe phase compensator 3112 bypasses the first cleaner 3113 and isinputted to the CIR interpolator-extrapolator 3114. Subsequently, thesecond cleaner 3115 cleans the CIR interpolated by the CIRinterpolator-extrapolator 3114.

Conversely, the CIR interpolator-extrapolator 3114 performs CIRextrapolation by using a difference value between two CIRs, each havingits phase change compensated by the phase compensator 3112, so as toestimate a CIR positioned outside of the two CIRs. Therefore, in thiscase, the noise element is rather amplified. Accordingly, when the CIRinterpolator-extrapolator 3114 performs CIR extrapolation, the CIRcleaned by the first cleaner 3113 is used. More specifically, when theCIR interpolator-extrapolator 3114 performs CIR extrapolation, theextrapolated CIR passes through the second cleaner 3115, thereby beinginputted to the zero-padding unit 3116.

Meanwhile, when a second frequency domain converter (or fast fouriertransform (FFT2)) 3121 converts the CIR, which has been cleaned andoutputted from the second cleaner 3115, to a frequency domain, thelength and of the inputted CIR and the FFT size may not match (or beidentical to one another). In other words, the CIR length may be smallerthan the FFT size. In this case, the zero-padding unit 3116 adds anumber of zeros ‘0’s corresponding to the difference between the FFTsize and the CIR length to the inputted CIR, thereby outputting theprocessed CIR to the second frequency domain converter (FFT2) 3121.Herein, the zero-padded CIR may correspond to one of the interpolatedCIR, extrapolated CIR, and the CIR estimated in the known data section.

The second frequency domain converter 3121 performs FFT on the CIR beingoutputted from the zero padding unit 3116, thereby converting the CIR toa frequency domain CIR. Then, the second frequency domain converter 3121outputs the converted CIR to the coefficient calculator 3122.

The coefficient calculator 3122 uses the frequency domain CIR beingoutputted from the second frequency domain converter 3121 to calculatethe equalization coefficient. Then, the coefficient calculator 3122outputs the calculated coefficient to the distortion compensator 3130.Herein, for example, the coefficient calculator 3122 calculates achannel equalization coefficient of the frequency domain that canprovide minimum mean square error (MMSE) from the CIR of the frequencydomain, which is outputted to the distortion compensator 3130.

The distortion compensator 3130 performs a complex number multiplicationon the overlapped data of the frequency domain being outputted from theFFT unit 3102 of the first frequency domain converter 3100 and theequalization coefficient calculated by the coefficient calculator 3122,thereby compensating the channel distortion of the overlapped data beingoutputted from the FFT unit 3102.

FIG. 57 illustrates a block diagram of a channel equalizer according toanother embodiment of the present invention. In other words, FIG. 57illustrates a block diagram showing another example of a channelequalizer by using different CIR estimation and application methods inaccordance with regions A, B, C, and D, when the data group is dividedinto the structure shown in FIG. 5.

More specifically, as shown in FIG. 5, known data that are sufficientlyare being periodically transmitted in regions A/B (i.e., MPH blocks B3to B8). Therefore, an indirect equalizing method using the CIR may beused herein. However, in regions C/D (i.e., MPH blocks B1, B2, B9, andB10), the known data are neither able to be transmitted at asufficiently long length nor able to be periodically and equallytransmitted. Therefore, it is inadequate to estimate the CIR by usingthe known data. Accordingly, in regions C/D, a direct equalizing methodin which an error is obtained from the output of the equalizer, so as toupdate the coefficient.

The examples presented in the embodiments of the present invention shownin FIG. 57 include a method of performing indirect channel equalizationby using a cyclic prefix on the data of regions A/B, and a method ofperforming direct channel equalization by using an overlap & save methodon the data of regions C/D.

Accordingly, referring to FIG. 57, the frequency domain channelequalizer includes a frequency domain converter 3510, a distortioncompensator 3520, a time domain converter 3530, a first coefficientcalculating unit 3540, a second coefficient calculating unit 3550, and acoefficient selector 3560.

Herein, the frequency domain converter 3510 includes an overlap unit3511, a select unit 3512, and a first FFT unit 3513.

The time domain converter 3530 includes an IFFT unit 3531, a save unit3532, and a select unit 3533.

The first coefficient calculating unit 3540 includes a CIR estimator3541, an average calculator 3542, and second FFT unit 3543, and acoefficient calculator 3544.

The second coefficient calculating unit 3550 includes a decision unit3551, a select unit 3552, a subtractor 3553, a zero-padding unit 3554, athird FFT unit 3555, a coefficient updater 3556, and a delay unit 3557.

Also, a multiplexer (MUX), which selects data that are currently beinginputted as the input data depending upon whether the data correspond toregions A/B or to regions C/D, may be used as the select unit 3512 ofthe frequency domain converter 3510, the select unit 3533 of the timedomain converter 3530, and the coefficient selector 3560.

In the channel equalizer having the above-described structure, as shownin FIG. 57, if the data being inputted correspond to the data of regionsA/B, the select unit 3512 of the frequency domain converter 3510 selectsthe input data and not the output data of the overlap unit 3511. In thesame case, the select unit 3533 of the time domain converter 3530selects the output data of the IFFT unit 3531 and not the output data ofthe save unit 3532. The coefficient selector 3560 selects theequalization coefficient being outputted from the first coefficientcalculating unit 3540.

Conversely, if the data being inputted correspond to the data of regionsC/D, the select unit 3512 of the frequency domain converter 3510 selectsthe output data of the overlap unit 3511 and not the input data. In thesame case, the select unit 3533 of the time domain converter 3530selects the output data of the save unit 3532 and not the output data ofthe IFFT unit 3531. The coefficient selector 3560 selects theequalization coefficient being outputted from the second coefficientcalculating unit 3550.

More specifically, the received data are inputted to the overlap unit3511 and select unit 3512 of the frequency domain converter 3510, and tothe first coefficient calculating unit 3540. If the inputted datacorrespond to the data of regions A/B, the select unit 3512 selects thereceived data, which are then outputted to the first FFT unit 3513. Onthe other hand, if the inputted data correspond to the data of regionsC/D, the select unit 3512 selects the data that are overlapped by theoverlap unit 3513 and are, then, outputted to the first FFT unit 3513.The first FFT unit 3513 performs FFT on the time domain data that areoutputted from the select unit 3512, thereby converting the time domaindata to frequency domain data. Then, the converted data are outputted tothe distortion compensator 3520 and the delay unit 3557 of the secondcoefficient calculating unit 3550.

The distortion compensator 3520 performs complex multiplication onfrequency domain data outputted from the first FFT unit 3513 and theequalization coefficient outputted from the coefficient selector 3560,thereby compensating the channel distortion detected in the data thatare being outputted from the first FFT unit 3513.

Thereafter, the distortion-compensated data are outputted to the IFFTunit 3531 of the time domain converter 3530. The IFFT unit 3531 of thetime domain converter 3530 performs IFFT on thechannel-distortion-compensated data, thereby converting the compensateddata to time domain data. The converted data are then outputted to thesave unit 3532 and the select unit 3533. If the inputted data correspondto the data of regions A/B, the select unit 3533 selects the output dataof the IFFT unit 3531. On the other hand, if the inputted datacorrespond to regions C/D, the select unit 3533 selects the valid dataextracted from the save unit 3532. Thereafter, the selected data areoutputted to be decoded and, simultaneously, outputted to the secondcoefficient calculating unit 3550.

The CIR estimator 3541 of the first coefficient calculating unit 3540uses the data being received during the known data section and the knowndata of the known data section, the known data being already known bythe receiving system in accordance with an agreement between thereceiving system and the transmitting system, in order to estimate theCIR. Subsequently, the estimated CIR is outputted to the averagecalculator 3542. The average calculator 3542 calculates an average valueof the CIRs that are being inputted consecutively. Then, the calculatedaverage value is outputted to the second FFT unit 3543. For example,referring to FIG. 37, the average value of the CIR value estimated atpoint T1 and the CIR value estimated at point T2 is used for the channelequalization process of the general data existing between point T1 andpoint T2. Accordingly, the calculated average value is outputted to thesecond FFT unit 3543.

The second FFT unit 3543 performs FFT on the CIR of the time domain thatis being inputted, so as to convert the inputted CIR to a frequencydomain CIR. Thereafter, the converted frequency domain CIR is outputtedto the coefficient calculator 3544. The coefficient calculator 3544calculates a frequency domain equalization coefficient that satisfiesthe condition of using the CIR of the frequency domain so as to minimizethe mean square error. The calculated equalizer coefficient of thefrequency domain is then outputted to the coefficient calculator 3560.

The decision unit 3551 of the second coefficient calculating unit 3550selects one of a plurality of decision values (e.g., 8 decision values)that is most approximate to the equalized data and outputs the selecteddecision value to the select unit 3552. Herein, a multiplexer may beused as the select unit 3552. In a general data section, the select unit3552 selects the decision value of the decision unit 3551.Alternatively, in a known data section, the select unit 3552 selects theknown data and outputs the selected known data to the subtractor 3553.The subtractor 3553 subtracts the output of the select unit 3533included in the time domain converter 3530 from the output of the selectunit 652 so as to calculate (or obtain) an error value. Thereafter, thecalculated error value is outputted to the zero-padding unit 3554.

The zero-padding unit 3554 adds (or inserts) the same amount of zeros(0) corresponding to the overlapped amount of the received data in theinputted error. Then, the error extended with zeros (0) is outputted tothe third FFT unit 3555. The third FFT unit 3555 converts the error ofthe time domain having zeros (0) added (or inserted) therein, to theerror of the frequency domain. Thereafter, the converted error isoutputted to the coefficient update unit 3556. The coefficient updateunit 3556 uses the received data of the frequency domain that have beendelayed by the delay unit 3557 and the error of the frequency domain soas to update the previous equalization coefficient. Thereafter, theupdated equalization coefficient is outputted to the coefficientselector 3560.

At this point, the updated equalization coefficient is stored so as thatit can be used as a previous equalization coefficient in a laterprocess. If the input data correspond to the data of regions A/B, thecoefficient selector 3560 selects the equalization coefficientcalculated from the first coefficient calculating unit 3540. On theother hand, if the input data correspond to the data of regions C/D, thecoefficient selector 3560 selects the equalization coefficient updatedby the second coefficient calculating unit 3550. Thereafter, theselected equalization coefficient is outputted to the distortioncompensator 3520.

FIG. 58 illustrates a block diagram of a channel equalizer according toanother embodiment of the present invention. In other words, FIG. 58illustrates a block diagram showing another example of a channelequalizer by using different CIR estimation and application methods inaccordance with regions A, B, C, and D, when the data group is dividedinto the structure shown in FIG. 5. In this example, a method ofperforming indirect channel equalization by using an overlap & savemethod on the data of regions A/B, and a method of performing directchannel equalization by using an overlap & save method on the data ofregions C/D are illustrated.

Accordingly, referring to FIG. 58, the frequency domain channelequalizer includes a frequency domain converter 3610, a distortioncompensator 3620, a time domain converter 3630, a first coefficientcalculating unit 3640, a second coefficient calculating unit 3650, and acoefficient selector 3660.

Herein, the frequency domain converter 3610 includes an overlap unit3611 and a first FFT unit 3612.

The time domain converter 3630 includes an IFFT unit 3631 and a saveunit 3632.

The first coefficient calculating unit 3640 includes a CIR estimator3641, an interpolator 3642, a second FFT unit 3643, and a coefficientcalculator 3644.

The second coefficient calculating unit 3650 includes a decision unit3651, a select unit 3652, a subtractor 3653, a zero-padding unit 3654, athird FFT unit 3655, a coefficient updater 3656, and a delay unit 3657.

Also, a multiplexer (MUX), which selects data that are currently beinginputted as the input data depending upon whether the data correspond toregions A/B or to regions C/D, may be used as the coefficient selector3660. More specifically, if the input data correspond to the data ofregions A/B, the coefficient selector 3660 selects the equalizationcoefficient calculated from the first coefficient calculating unit 3640.On the other hand, if the input data correspond to the data of regionsC/D, the coefficient selector 3660 selects the equalization coefficientupdated by the second coefficient calculating unit 3650.

In the channel equalizer having the above-described structure, as shownin FIG. 58, the received data are inputted to the overlap unit 3611 ofthe frequency domain converter 3610 and to the first coefficientcalculating unit 3640. The overlap unit 3611 overlaps the input data toa pre-determined overlapping ratio and outputs the overlapped data tothe first FFT unit 3612. The first FFT unit 3612 performs FFT on theoverlapped time domain data, thereby converting the overlapped timedomain data to overlapped frequency domain data. Then, the converteddata are outputted to the distortion compensator 3620 and the delay unit3657 of the second coefficient calculating unit 3650.

The distortion compensator 3620 performs complex multiplication on theoverlapped frequency domain data outputted from the first FFT unit 3612and the equalization coefficient outputted from the coefficient selector3660, thereby compensating the channel distortion detected in theoverlapped data that are being outputted from the first FFT unit 3612.Thereafter, the distortion-compensated data are outputted to the IFFTunit 3631 of the time domain converter 3630. The IFFT unit 3631 of thetime domain converter 3630 performs IFFT on the distortion-compensateddata, thereby converting the compensated data to overlapped time domaindata. The converted overlapped data are then outputted to the save unit3632. The save unit 3632 extracts only the valid data from theoverlapped time domain data, which are then outputted for data decodingand, at the same time, outputted to the second coefficient calculatingunit 3650 in order to update the coefficient.

The CIR estimator 3641 of the first coefficient calculating unit 3640uses the data received during the known data section and the known datain order to estimate the CIR. Subsequently, the estimated CIR isoutputted to the interpolator 3642. The interpolator 3642 uses theinputted CIR to estimate the CIRs (i.e., CIRs of the region that doesnot include the known data) corresponding to the points located betweenthe estimated CIRs according to a predetermined interpolation method.Thereafter, the estimated result is outputted to the second FFT unit3643. The second FFT unit 3643 performs FFT on the inputted CIR, so asto convert the inputted CIR to a frequency domain CIR. Thereafter, theconverted frequency domain CIR is outputted to the coefficientcalculator 3644. The coefficient calculator 3644 calculates a frequencydomain equalization coefficient that satisfies the condition of usingthe CIR of the frequency domain so as to minimize the mean square error.The calculated equalizer coefficient of the frequency domain is thenoutputted to the coefficient calculator 3660.

The structure and operations of the second coefficient calculating unit3650 is identical to those of the second coefficient calculating unit3550 shown in FIG. 57. Therefore, the description of the same will beomitted for simplicity.

If the input data correspond to the data of regions A/B, the coefficientselector 3660 selects the equalization coefficient calculated from thefirst coefficient calculating unit 3640. On the other hand, if the inputdata correspond to the data of regions C/D, the coefficient selector3660 selects the equalization coefficient updated by the secondcoefficient calculating unit 3650. Thereafter, the selected equalizationcoefficient is outputted to the distortion compensator 3620.

FIG. 59 illustrates a block diagram of a channel equalizer according toanother embodiment of the present invention. In other words, FIG. 59illustrates a block diagram showing another example of a channelequalizer by using different CIR estimation and application methods inaccordance with regions A, B, C, and D, when the data group is dividedinto the structure shown in FIG. 5. For example, in regions A/B, thepresent invention uses the known data in order to estimate the CIR byusing a least square (LS) method, thereby performing the channelequalization process. On the other hand, in regions C/D, the presentinvention estimates the CIR by using a least mean square (LMS) method,thereby performing the channel equalization process. More specifically,since the periodic known data do not exist in regions C/D, as in regionsA/B, the same channel equalization process as that of regions A/B cannotbe performed in regions C/D. Therefore, the channel equalization processmay only be performed by using the LMS method.

Referring to FIG. 59, the channel equalizer includes an overlap unit3701, a first fast fourier transform (FFT) unit 3702, a distortioncompensator 3703, an inverse fast fourier transform (IFFT) unit 3704, asave unit 3705, a first CIR estimator 3706, a CIR interpolator 3707, adecision unit 3708, a second CIR estimator 3710, a selection unit 3711,a second FFT unit 3712, and a coefficient calculator 3713. Herein, anydevice performed complex number multiplication may be used as thedistortion compensator 3703. In the channel equalizer having theabove-described structure, as shown in FIG. 59, the overlap unit 3701overlaps the data being inputted to the channel equalizer to apredetermined overlapping ratio and then outputs the overlapped data tothe first FFT unit 3702. The first FFT unit 3702 converts (ortransforms) the overlapped data of the time domain to overlapped data ofthe frequency domain by using fast fourier transform (FFT). Then, theconverted data are outputted to the distortion compensator 3703.

The distortion converter 3703 performs complex multiplication on theequalization coefficient calculated from the coefficient calculator 3713and the overlapped data of the frequency domain, thereby compensatingthe channel distortion of the overlapped data being outputted from thefirst FFT unit 3702. Thereafter, the distortion-compensated data areoutputted to the IFFT unit 3704. The IFFT unit 3704 performs inversefast fourier transform (IFFT) on the distortion-compensated overlappeddata, so as to convert the corresponding data back to data (i.e.,overlapped data) of the time domain. Subsequently, the converted dataare outputted to the save unit 3705. The save unit 3705 extracts onlythe valid data from the overlapped data of the time domain. Then, thesave unit 3705 outputs the extracted valid data for a data decodingprocess and, at the same time, outputs the extracted valid data to thedecision unit 3708 for a channel estimation process.

The decision unit 3708 selects one of a plurality of decision values(e.g., 8 decision values) that is most approximate to the equalized dataand outputs the selected decision value to the select unit 3709. Herein,a multiplexer may be used as the select unit 3709. In a general datasection, the select unit 3709 selects the decision value of the decisionunit 3708. Alternatively, in a known data section, the select unit 3709selects the known data and outputs the selected known data to the secondCIR estimator 3710.

Meanwhile, the first CIR estimator 3706 uses the data that are beinginputted in the known data section and the known data so as to estimatethe CIR.

Thereafter, the first CIR estimator 3706 outputs the estimated CIR tothe CIR interpolator 3707. Herein, the known data correspond toreference known data created during the known data section by thereceiving system in accordance to an agreement between the transmittingsystem and the receiving system. At this point, according to anembodiment of the present invention, the first CIR estimator 3706 usesthe LS method to estimate the CIR. The LS estimation method calculates across correlation value p between the known data that have passedthrough the channel during the known data section and the known datathat are already known by the receiving end. Then, a cross correlationmatrix R of the known data is calculated. Subsequently, a matrixoperation is performed on R⁻¹·p so that the cross correlation portionwithin the cross correlation value p between the received data and theinitial known data, thereby estimating the CIR of the transmissionchannel.

The CIR interpolator 3707 receives the CIR from the first CIR estimator3706. And, in the section between two sets of known data, the CIR isinterpolated in accordance with a pre-determined interpolation method.Then, the interpolated CIR is outputted. At this point, thepre-determined interpolation method corresponds to a method ofestimating a particular set of data at an unknown point by using a setof data known by a particular function. For example, such methodincludes a linear interpolation method. The linear interpolation methodis only one of the most simple interpolation methods. A variety of otherinterpolation methods may be used instead of the above-described linearinterpolation method. It is apparent that the present invention is notlimited only to the example set forth in the description of the presentinvention. More specifically, the CIR interpolator 3707 uses the CIRthat is being inputted in order to estimate the CIR of the section thatdoes not include any known data by using the pre-determinedinterpolation method. Thereafter, the estimated CIR is outputted to theselect unit 3711.

The second CIR estimator 3710 uses the input data of the channelequalizer and the output data of the select unit 3709 in order toestimate the CIR. Then, the second CIR estimator 3710 outputs theestimated CIR to the select unit 3711. At this point, according to anembodiment of the present invention, the CIR is estimated by using theLMS method. The LMS estimation method will be described in detail in alater process.

In regions A/B (i.e., MPH blocks B3 to B8), the select unit 3711 selectsthe CIR outputted from the CIR interpolator 3707. And, in regions C/D(i.e., MPH blocks B1, B2, B9, and B10), the select unit 3711 selects theCIR outputted from the second CIR estimator 3710. Thereafter, the selectunit 3711 outputs the selected CIR to the second FFT unit 3712.

The second FFT unit 3712 converts the CIR that is being inputted to aCIR of the frequency domain, which is then outputted to the coefficientcalculator 3713. The coefficient calculator 3713 uses the CIR of thefrequency domain that is being inputted, so as to calculate theequalization coefficient and to output the calculated equalizationcoefficient to the distortion compensator 3703. At this point, thecoefficient calculator 3713 calculates a channel equalizationcoefficient of the frequency domain that can provide minimum mean squareerror (MMSE) from the CIR of the frequency domain. At this point, thesecond CIR estimator 3710 may use the CIR estimated in regions A/B asthe CIR at the beginning of regions C/D. For example, the CIR value ofMPH block B8 may be used as the CIR value at the beginning of the MPHblock B9. Accordingly, the convergence speed of regions C/D may bereduced.

The basic principle of estimating the CIR by using the LMS method in thesecond CIR estimator 3710 corresponds to receiving the output of anunknown transmission channel and to updating (or renewing) thecoefficient of an adaptive filter (not shown) so that the differencevalue between the output value of the unknown channel and the outputvalue of the adaptive filter is minimized. More specifically, thecoefficient value of the adaptive filter is renewed so that the inputdata of the channel equalizer is equal to the output value of theadaptive filter (not shown) included in the second CIR estimator 3710.Thereafter, the filter coefficient is outputted as the CIR after eachFFT cycle.

Referring to FIG. 60, the second CIR estimator 3710 includes a delayunit T, a multiplier, and a coefficient renewal unit for each tab.Herein, the delay unit T sequentially delays the output data x(n) of theselect unit 3709. The multiplier multiplies respective output dataoutputted from each delay unit T with error data e(n). The coefficientrenewal unit renews the coefficient by using the output corresponding toeach multiplier. Herein, the multipliers that are being provided as manyas the number of tabs will be referred to as a first multiplying unitfor simplicity. Furthermore, the second CIR estimator 3710 furtherincludes a plurality of multipliers each multiplying the output data ofthe select unit 3709 and the output data of the delay unit T (whereinthe output data of the last delay unit are excluded) with the outputdata corresponding to each respective coefficient renewal unit. Thesemultipliers are also provided as many as the number of tabs. This groupof multipliers will be referred to as a second multiplying unit forsimplicity.

The second CIR estimator 3710 further includes an adder and asubtractor. Herein, the adder adds all of the data outputted from eachmultipliers included in the second multiplier unit. Then, the addedvalue is outputted as the estimation value ŷ(n) of the data inputted tothe channel equalizer. The subtractor calculates the difference betweenthe output data ŷ(n) of the adder and the input data y(n) of the channelequalizer. Thereafter, the calculated difference value is outputted asthe error data e(n). Referring to FIG. 60, in a general data section,the decision value of the equalized data is inputted to the first delayunit included in the second CIR estimator 3710 and to the firstmultiplier included in the second multiplier. In the known data section,the known data are inputted to the first delay unit included in thesecond CIR estimator 3710 and to the first multiplier included in thesecond multiplier unit. The input data {circumflex over (x)}(n) aresequentially delayed by passing through a number of serially connecteddelay units T, the number corresponding to the number of tabs. Theoutput data of each delay unit T and the error data e(n) are multipliedby each corresponding multiplier included in the first multiplier unit.Thereafter, the coefficients are renewed by each respective coefficientrenewal unit.

Each coefficient that is renewed by the corresponding coefficientrenewal unit is multiplied with the input data the output data{circumflex over (x)}(n) and also with the output data of each delayunit T with the exception of the last delay. Thereafter, the multipliedvalue is inputted to the adder. The adder then adds all of the outputdata outputted from the second multiplier unit and outputs the addedvalue to the subtractor as the estimation value ŷ(n) of the input dataof the channel equalizer. The subtractor calculates a difference valuebetween the estimation value ŷ(n) and the input data y(n) of the channelequalizer. The difference value is then outputted to each multiplier ofthe first multiplier unit as the error data e(n). At this point, theerror data e(n) is outputted to each multiplier of the first multiplierunit by passing through each respective delay unit T. As describedabove, the coefficient of the adaptive filter is continuously renewed.And, the output of each coefficient renewal unit is outputted as the CIRof the second CIR estimator 3710 after each FFT cycle.

Block Decoder

Meanwhile, if the data being inputted to the block decoder 1005, afterbeing channel-equalized by the equalizer 1003, correspond to the datahaving both block encoding and trellis encoding performed thereon (i.e.,the data within the RS frame, the signaling information data, etc.) bythe transmitting system, trellis decoding and block decoding processesare performed on the inputted data as inverse processes of thetransmitting system. Alternatively, if the data being inputted to theblock decoder 1005 correspond to the data having only trellis encodingperformed thereon (i.e., the main service data), and not the blockencoding, only the trellis decoding process is performed on the inputteddata as the inverse process of the transmitting system.

The trellis decoded and block decoded data by the block decoder 1005 arethen outputted to the RS frame decoder 1006. More specifically, theblock decoder 1005 removes the known data, data used for trellisinitialization, and signaling information data, MPEG header, which havebeen inserted in the data group, and the RS parity data, which have beenadded by the RS encoder/non-systematic RS encoder or non-systematic RSencoder of the transmitting system. Then, the block decoder 1005 outputsthe processed data to the RS frame decoder 1006. Herein, the removal ofthe data may be performed before the block decoding process, or may beperformed during or after the block decoding process.

Meanwhile, the data trellis-decoded by the block decoder 1005 areoutputted to the data deinterleaver 1009. At this point, the data beingtrellis-decoded by the block decoder 1005 and outputted to the datadeinterleaver 1009 may not only include the main service data but mayalso include the data within the RS frame and the signaling information.Furthermore, the RS parity data that are added by the transmittingsystem after the pre-processor 230 may also be included in the databeing outputted to the data deinterleaver 1009.

According to another embodiment of the present invention, data that arenot processed with block decoding and only processed with trellisencoding by the transmitting system may directly bypass the blockdecoder 1005 so as to be outputted to the data deinterleaver 1009. Inthis case, a trellis decoder should be provided before the datadeinterleaver 1009. More specifically, if the inputted data correspondto the data having only trellis encoding performed thereon and not blockencoding, the block decoder 1005 performs Viterbi (or trellis) decodingon the inputted data so as to output a hard decision value or to performa hard-decision on a soft decision value, thereby outputting the result.

Meanwhile, if the inputted data correspond to the data having both blockencoding process and trellis encoding process performed thereon, theblock decoder 1005 outputs a soft decision value with respect to theinputted data.

In other words, if the inputted data correspond to data being processedwith block encoding by the block processor 302 and being processed withtrellis encoding by the trellis encoding module 256, in the transmittingsystem, the block decoder 1005 performs a decoding process and a trellisdecoding process on the inputted data as inverse processes of thetransmitting system. At this point, the RS frame encoder of thepre-processor included in the transmitting system may be viewed as anouter (or external) encoder. And, the trellis encoder may be viewed asan inner (or internal) encoder. When decoding such concatenated codes,in order to allow the block decoder 1005 to maximize its performance ofdecoding externally encoded data, the decoder of the internal codeshould output a soft decision value.

FIG. 61 illustrates a detailed block diagram of the block decoder 1005according to an embodiment of the present invention. Referring to FIG.61, the block decoder 1005 includes a feedback controller 4010, an inputbuffer 4011, a trellis decoding unit (or 12-way trellis coded modulation(TCM) decoder or inner decoder) 4012, a symbol-byte converter 4013, anouter block extractor 4014, a feedback deformatter 4015, a symboldeinterleaver 4016, an outer symbol mapper 4017, a symbol decoder 4018,an inner symbol mapper 4019, a symbol interleaver 4020, a feedbackformatter 4021, and an output buffer 4022. Herein, just as in thetransmitting system, the trellis decoding unit 4012 may be viewed as aninner (or internal) decoder. And, the symbol decoder 4018 may be viewedas an outer (or external) decoder.

The input buffer 4011 temporarily stores the mobile service data symbolsbeing channel-equalized and outputted from the equalizer 1003. (Herein,the mobile service data symbols may include symbols corresponding to thesignaling information, RS parity data symbols and CRC data symbols addedduring the encoding process of the RS frame.) Thereafter, the inputbuffer 4011 repeatedly outputs the stored symbols for M number of timesto the trellis decoding unit 4012 in a turbo block (TDL) size requiredfor the turbo decoding process.

The turbo decoding length (TDL) may also be referred to as a turboblock. Herein, a TDL should include at least one SCCCC block size.Therefore, as defined in FIG. 5, when it is assumed that one MPH blockis a 16-segment unit, and that a combination of 10 MPH blocks form oneSCCC block, a TDL should be equal to or larger than the maximum possiblecombination size. For example, when it is assumed that 2 MPH blocks formone SCCC block, the TDL may be equal to or larger than 32 segments(i.e., 828×32=26496 symbols). Herein, M indicates a number ofrepetitions for turbo-decoding pre-decided by the feed-back controller4010.

Also, M represents a number of repetitions of the turbo decodingprocess, the number being predetermined by the feedback controller 4010.

Additionally, among the values of symbols being channel-equalized andoutputted from the equalizer 1003, the input symbol values correspondingto a section having no mobile service data symbols (including RS paritydata symbols during RS frame encoding and CRC data symbols) includedtherein, bypass the input buffer 4011 without being stored. Morespecifically, since trellis-encoding is performed on input symbol valuesof a section wherein SCCC block-encoding has not been performed, theinput buffer 4011 inputs the inputted symbol values of the correspondingsection directly to the trellis encoding module 4012 without performingany storage, repetition, and output processes. The storage, repetition,and output processes of the input buffer 4011 are controlled by thefeedback controller 4010. Herein, the feedback controller 4010 refers toSCCC-associated information (e.g., SCCC block mode and SCCC outer codemode), which are outputted from the signaling information decoding unit1013, in order to control the storage and output processes of the inputbuffer 4011.

The trellis decoding unit 4012 includes a 12-way TCM decoder. Herein,the trellis decoding unit 4012 performs 12-way trellis decoding asinverse processes of the 12-way trellis encoder.

More specifically, the trellis decoding unit 4012 receives a number ofoutput symbols of the input buffer 4011 and soft-decision values of thefeedback formatter 4021 equivalent to each TDL, so as to perform the TCMdecoding process.

At this point, based upon the control of the feedback controller 4010,the soft-decision values outputted from the feedback formatter 4021 arematched with a number of mobile service data symbol places so as to bein a one-to-one (1:1) correspondence. Herein, the number of mobileservice data symbol places is equivalent to the TDL being outputted fromthe input buffer 4011.

More specifically, the mobile service data being outputted from theinput buffer 4011 are matched with the turbo decoded data beinginputted, so that each respective data place can correspond with oneanother. Thereafter, the matched data are outputted to the trellisdecoding unit 4012. For example, if the turbo decoded data correspond tothe third symbol within the turbo block, the corresponding symbol (ordata) is matched with the third symbol included in the turbo block,which is outputted from the input buffer 4011. Subsequently, the matchedsymbol (or data) is outputted to the trellis decoding unit 4012.

In order to do so, while the regressive turbo decoding is in process,the feedback controller 4010 controls the input buffer 4011 so that theinput buffer 4011 stores the corresponding turbo block data. Also, bydelaying data (or symbols), the soft decision value (e.g., LLR) of thesymbol outputted from the symbol interleaver 4020 and the symbol of theinput buffer 4011 corresponding to the same place (or position) withinthe block of the output symbol are matched with one another to be in aone-to-one correspondence. Thereafter, the matched symbols arecontrolled so that they can be inputted to the TCM decoder through therespective path. This process is repeated for a predetermined number ofturbo decoding cycle periods. Then, the data of the next turbo block areoutputted from the input buffer 4011, thereby repeating the turbodecoding process.

The output of the trellis decoding unit 4012 signifies a degree ofreliability of the transmission bits configuring each symbol. Forexample, in the transmitting system, since the input data of the trellisencoding module correspond to two bits as one symbol, a log likelihoodratio (LLR) between the likelihood of a bit having the value of ‘1’ andthe likelihood of the bit having the value of ‘0’ may be respectivelyoutputted (in bit units) to the upper bit and the lower bit. Herein, thelog likelihood ratio corresponds to a log value for the ratio betweenthe likelihood of a bit having the value of ‘1’ and the likelihood ofthe bit having the value of ‘0’. Alternatively, a LLR for the likelihoodof 2 bits (i.e., one symbol) being equal to “00”, “01”, “10”, and “11”may be respectively outputted (in symbol units) to all 4 combinations ofbits (i.e., 00, 01, 10, 11). Consequently, this becomes the softdecision value that indicates the degree of reliability of thetransmission bits configuring each symbol. A maximum a posterioriprobability (MAP) or a soft-out Viterbi algorithm (SOVA) may be used asa decoding algorithm of each TCM decoder within the trellis decodingunit 4012.

The output of the trellis decoding unit 4012 is inputted to thesymbol-byte converter 4013 and the outer block extractor 4014.

The symbol-byte converter 4013 performs a hard-decision process of thesoft decision value that is trellis decoded and outputted from thetrellis decoding unit 4012. Thereafter, the symbol-byte converter 4013groups 4 symbols into byte units, which are then outputted to the datadeinterleaver 1009 of FIG. 36. More specifically, the symbol-byteconverter 4013 performs hard-decision in bit units on the soft decisionvalue of the symbol outputted from the trellis decoding unit 4012.Therefore, the data processed with hard-decision and outputted in bitunits from the symbol-byte converter 4013 not only include main servicedata, but may also include mobile service data, known data, RS paritydata, and MPEG headers.

Among the soft decision values of TDL size of the trellis decoding unit4012, the outer block extractor 4014 identifies the soft decision valuesof B size of corresponding to the mobile service data symbols (whereinsymbols corresponding to signaling information, RS parity data symbolsthat are added during the encoding of the RS frame, and CRC data symbolsare included) and outputs the identified soft decision values to thefeedback deformatter 4015.

The feedback deformatter 4015 changes the processing order of the softdecision values corresponding to the mobile service data symbols. Thisis an inverse process of an initial change in the processing order ofthe mobile service data symbols, which are generated during anintermediate step, wherein the output symbols outputted from the blockprocessor 302 of the transmitting system are being inputted to thetrellis encoding module 256 (e.g., when the symbols pass through thegroup formatter, the data deinterleaver, the packet formatter, and thedata interleaver). Thereafter, the feedback deformatter 1015 performsreordering of the process order of soft decision values corresponding tothe mobile service data symbols and, then, outputs the processed mobileservice data symbols to the symbol deinterleaver 4016.

This is because a plurality of blocks exist between the block processor302 and the trellis encoding module 256, and because, due to theseblocks, the order of the mobile service data symbols being outputtedfrom the block processor 302 and the order of the mobile service datasymbols being inputted to the trellis encoding module 256 are notidentical to one another. Therefore, the feedback deformatter 4015reorders (or rearranges) the order of the mobile service data symbolsbeing outputted from the outer block extractor 4014, so that the orderof the mobile service data symbols being inputted to the symboldeinterleaver 4016 matches the order of the mobile service data symbolsoutputted from the block processor 302 of the transmitting system. Thereordering process may be embodied as one of software, middleware, andhardware.

FIG. 62 illustrates a detailed block view of the feedback deformatter4015 according to an embodiment of the present invention. Herein, thefeedback deformatter 4015 includes a data deinterleaver 5011, a packetdeformatter 5012, a data interleaver 5013, and a group deformatter 5014.Referring to FIG. 62, the soft decision value of the mobile service datasymbol, which is extracted by the outer block extractor 4014, isoutputted directly to the data deinterleaver 5011 of the feedbackdeformatter 4015 without modification. However, data place holders (ornull data) are inserted in data places (e.g., main service data places,known data places, signaling information places, RS parity data places,and MPEG header places), which are removed by the outer block extractor4014, thereby being outputted to the data deinterleaver 5011 of thefeedback deformatter 4015.

The data deinterleaver 5011 performs an inverse process of the datainterleaver 253 included in the transmitting system. More specifically,the data deinterleaver 5011 deinterleaves the inputted data and outputsthe deinterleaved data to the packet deformatter 5012. The packetdeformatter 5012 performs an inverse process of the packet formatter305. More specifically, among the data that are deinterleaved andoutputted from the data deinterleaver 5011, the packet deformatter 5012removes the place holder corresponding to the MPEG header, which hadbeen inserted to the packet formatter 305. The output of the packetdeformatter 5012 is inputted to the data interleaver 5013, and the datainterleaver 5013 interleaves the data being inputted, as an inverseprocess of the data deinterleaver 529 included in the transmittingsystem. Accordingly, data having a data structure as shown in FIG. 5,are outputted to the group deformatter 5014.

The data deformatter 5014 performs an inverse process of the groupformatter 303 included in the transmitting system. More specifically,the group formatter 5014 removes the place holders corresponding to themain service data, known data, signaling information data, and RS paritydata. Then, the group formatter 5014 outputs only the reordered (orrearranged) mobile service data symbols to the symbol deinterleaver4016. According to another embodiment of the present invention, when thefeedback deformatter 4015 is embodied using a memory map, the process ofinserting and removing place holder to and from data places removed bythe outer block extractor 4014 may be omitted.

The symbol deinterleaver 4016 performs deinterleaving on the mobileservice data symbols having their processing orders changed andoutputted from the feedback deformatter 4015, as an inverse process ofthe symbol interleaving process of the symbol interleaver 514 includedin the transmitting system. The size of the block used by the symboldeinterleaver 4016 during the deinterleaving process is identical tointerleaving size of an actual symbol (i.e., B) of the symbolinterleaver 514, which is included in the transmitting system. This isbecause the turbo decoding process is performed between the trellisdecoding unit 4012 and the symbol decoder 4018. Both the input andoutput of the symbol deinterleaver 4016 correspond to soft decisionvalues, and the deinterleaved soft decision values are outputted to theouter symbol mapper 4017.

The operations of the outer symbol mapper 4017 may vary depending uponthe structure and coding rate of the convolution encoder 513 included inthe transmitting system. For example, when data are ½-rate encoded bythe convolution encoder 513 and then transmitted, the outer symbolmapper 4017 directly outputs the input data without modification. Inanother example, when data are ¼-rate encoded by the convolution encoder513 and then transmitted, the outer symbol mapper 4017 converts theinput data so that it can match the input data format of the symboldecoder 4018. For this, the outer symbol mapper 4017 may be inputtedSCCC-associated information (i.e., SCCC block mode and SCCC outer codemode) from the signaling information decoder 1013. Then, the outersymbol mapper 4017 outputs the converted data to the symbol decoder4018.

The symbol decoder 4018 (i.e., the outer decoder) receives the dataoutputted from the outer symbol mapper 4017 and performs symbol decodingas an inverse process of the convolution encoder 513 included in thetransmitting system. At this point, two different soft decision valuesare outputted from the symbol decoder 4018. One of the outputted softdecision values corresponds to a soft decision value matching the outputsymbol of the convolution encoder 513 (hereinafter referred to as a“first decision value”). The other one of the outputted soft decisionvalues corresponds to a soft decision value matching the input bit ofthe convolution encoder 513 (hereinafter referred to as a “seconddecision value”).

More specifically, the first decision value represents a degree ofreliability the output symbol (i.e., 2 bits) of the convolution encoder513. Herein, the first soft decision value may output (in bit units) aLLR between the likelihood of 1 bit being equal to ‘1’ and thelikelihood of 1 bit being equal to ‘0’ with respect to each of the upperbit and lower bit, which configures a symbol. Alternatively, the firstsoft decision value may also output (in symbol units) a LLR for thelikelihood of 2 bits being equal to “00”, “01”, “10”, and “11” withrespect to all possible combinations. The first soft decision value isfed-back to the trellis decoding unit 4012 through the inner symbolmapper 4019, the symbol interleaver 4020, and the feedback formatter4021. On the other hand, the second soft decision value indicates adegree of reliability the input bit of the convolution encoder 513included in the transmitting system. Herein, the second soft decisionvalue is represented as the LLR between the likelihood of 1 bit beingequal to ‘1’ and the likelihood of 1 bit being equal to ‘0’. Thereafter,the second soft decision value is outputted to the outer buffer 4022. Inthis case, a maximum a posteriori probability (MAP) or a soft-outViterbi algorithm (SOVA) may be used as the decoding algorithm of thesymbol decoder 4018.

The first soft decision value that is outputted from the symbol decoder4018 is inputted to the inner symbol mapper 4019. The inner symbolmapper 4019 converts the first soft decision value to a data formatcorresponding the input data of the trellis decoding unit 4012.Thereafter, the inner symbol mapper 4019 outputs the converted softdecision value to the symbol interleaver 4020. The operations of theinner symbol mapper 4019 may also vary depending upon the structure andcoding rate of the convolution encoder 513 included in the transmittingsystem.

The symbol interleaver 4020 performs symbol interleaving, as shown inFIG. 26, on the first soft decision value that is outputted from theinner symbol mapper 4019. Then, the symbol interleaver 4020 outputs thesymbol-interleaved first soft decision value to the feedback formatter4021. Herein, the output of the symbol interleaver 4020 also correspondsto a soft decision value.

With respect to the changed processing order of the soft decision valuescorresponding to the symbols that are generated during an intermediatestep, wherein the output symbols outputted from the block processor 302of the transmitting system are being inputted to the trellis encodingmodule (e.g., when the symbols pass through the group formatter, thedata deinterleaver, the packet formatter, the RS encoder, and the datainterleaver), the feedback formatter 4021 alters (or changes) the orderof the output values outputted from the symbol interleaver 4020.Subsequently, the feedback formatter 4020 outputs values to the trellisdecoding unit 4012 in the changed order. The reordering process of thefeedback formatter 4021 may configure at least one of software,hardware, and middleware. For example, the feedback formatter 4021 mayconfigure to be performed as an inverse process of FIG. 62.

The soft decision values outputted from the symbol interleaver 4020 arematched with the positions of mobile service data symbols each havingthe size of TDL, which are outputted from the input buffer 4011, so asto be in a one-to-one correspondence. Thereafter, the soft decisionvalues matched with the respective symbol position are inputted to thetrellis decoding unit 4012. At this point, since the main service datasymbols or the RS parity data symbols and known data symbols of the mainservice data do not correspond to the mobile service data symbols, thefeedback formatter 4021 inserts null data in the correspondingpositions, thereby outputting the processed data to the trellis decodingunit 4012. Additionally, each time the symbols having the size of TDLare turbo decoded, no value is fed-back by the symbol interleaver 4020starting from the beginning of the first decoding process. Therefore,the feedback formatter 4021 is controlled by the feedback controller4010, thereby inserting null data into all symbol positions including amobile service data symbol. Then, the processed data are outputted tothe trellis decoding unit 4012.

The output buffer 4022 receives the second soft decision value from thesymbol decoder 4018 based upon the control of the feedback controller4010. Then, the output buffer 4022 temporarily stores the receivedsecond soft decision value. Thereafter, the output buffer 4022 outputsthe second soft decision value to the RS frame decoder 10006. Forexample, the output buffer 4022 overwrites the second soft decisionvalue of the symbol decoder 4018 until the turbo decoding process isperformed for M number of times. Then, once all M number of turbodecoding processes is performed for a single TDL, the correspondingsecond soft decision value is outputted to the RS frame decoder 1006.

The feedback controller 4010 controls the number of turbo decoding andturbo decoding repetition processes of the overall block decoder, shownin FIG. 61. More specifically, once the turbo decoding process has beenrepeated for a predetermined number of times, the second soft decisionvalue of the symbol decoder 4018 is outputted to the RS frame decoder1006 through the output buffer 4022. Thus, the block decoding process ofa turbo block is completed. In the description of the present invention,this process is referred to as a regressive turbo decoding process forsimplicity.

At this point, the number of regressive turbo decoding rounds betweenthe trellis decoding unit 4012 and the symbol decoder 4018 may bedefined while taking into account hardware complexity and errorcorrection performance. Accordingly, if the number of rounds increases,the error correction performance may be enhanced. However, this may leadto a disadvantageous of the hardware becoming more complicated (orcomplex).

Meanwhile, the data deinterleaver 1009, the RS decoder 1010, and thedata derandomizer 1011 correspond to blocks required for receiving themain service data. Therefore, the above-mentioned blocks may not benecessary (or required) in the structure of a digital broadcastreceiving system for receiving mobile service data only.

The data deinterleaver 1009 performs an inverse process of the datainterleaver included in the transmitting system. In other words, thedata deinterleaver 1009 deinterleaves the main service data outputtedfrom the block decoder 1005 and outputs the deinterleaved main servicedata to the RS decoder 1010. The data being inputted to the datadeinterleaver 1009 include main service data, as well as mobile servicedata, known data, RS parity data, and an MPEG header. At this point,among the inputted data, only the main service data and the RS paritydata added to the main service data packet may be outputted to the RSdecoder 1010. Also, all data outputted after the data derandomizer 1011may all be removed with the exception for the main service data. In theembodiment of the present invention, only the main service data and theRS parity data added to the main service data packet are inputted to theRS decoder 1010.

The RS decoder 1010 performs a systematic RS decoding process on thedeinterleaved data and outputs the processed data to the dataderandomizer 1011.

The data derandomizer 1011 receives the output of the RS decoder 1010and generates a pseudo random data byte identical to that of therandomizer included in the digital broadcast transmitting system.Thereafter, the data derandomizer 1011 performs a bitwise exclusive OR(XOR) operation on the generated pseudo random data byte, therebyinserting the MPEG synchronization bytes to the beginning of each packetso as to output the data in 188-byte main service data packet units.

RS Frame Decoder

The data outputted from the block decoder 1005 are in portion units.More specifically, in the transmitting system, the RS frame is dividedinto several portions, and the mobile service data of each portion areassigned either to regions A/B/C/D within the data group or to any oneof regions A/B and regions C/D, thereby being transmitted to thereceiving system. Therefore, the RS frame decoder 1006 groups severalportions included in a parade so as to form an RS frame. Alternatively,the RS frame decoder 1006 may also group several portions included in aparade so as to form two RS frames. Thereafter, error correctiondecoding is performed in RS frame units.

For example, when the RS frame mode value is equal to ‘00’, then oneparade transmits one RS frame. At this point, one RS frame is dividedinto several portions, and the mobile service data of each portion areassigned to regions A/B/C/D of the corresponding data group, therebybeing transmitted. In this case, the MPH frame decoder 1006 extractsmobile service data from regions A/B/C/D of the corresponding datagroup, as shown in FIG. 63( a). Subsequently, the MPH frame decoder 1006may perform the process of forming (or creating) a portion on aplurality of data group within a parade, thereby forming severalportions. Then, the several portions of mobile service data may begrouped to form an RS frame. Herein, if stuffing bytes are added to thelast portion, the RS frame may be formed after removing the stuffingbyte.

In another example, when the RS frame mode value is equal to ‘01’, thenone parade transmits two RS frames (i.e., a primary RS frame and asecondary RS frame). At this point, a primary RS frame is divided intoseveral primary portions, and the mobile service data of each primaryportion are assigned to regions A/B of the corresponding data group,thereby being transmitted. Also, a secondary RS frame is divided intoseveral secondary portions, and the mobile service data of eachsecondary portion are assigned to regions C/D of the corresponding datagroup, thereby being transmitted.

In this case, the MPH frame decoder 1006 extracts mobile service datafrom regions A/B of the corresponding data group, as shown in FIG. 63(b). Subsequently, the MPH frame decoder 1006 may perform the process offorming (or creating) a primary portion on a plurality of data groupwithin a parade, thereby forming several primary portions. Then, theseveral primary portions of mobile service data may be grouped to form aprimary RS frame. Herein, if stuffing bytes are added to the lastprimary portion, the primary RS frame may be formed after removing thestuffing byte. Also, the MPH frame decoder 1006 extracts mobile servicedata from regions C/D of the corresponding data group. Subsequently, theMPH frame decoder 1006 may perform the process of forming (or creating)a secondary portion on a plurality of data group within a parade,thereby forming several secondary portions. Then, the several secondaryportions of mobile service data may be grouped to form a secondary RSframe. Herein, if stuffing bytes are added to the last secondaryportion, the secondary RS frame may be formed after removing thestuffing byte.

More specifically, the RS frame decoder 1006 receives the RS-encodedand/or CRC-encoded mobile service data of each portion from the blockdecoder 1005. Then, the RS frame decoder 1006 groups several portions,which are inputted based upon RS frame-associated information outputtedfrom the signaling information decoder 1013, thereby performing errorcorrection. By referring to the RS frame mode value included in the RSframe-associated information, the RS frame decoder 1006 may form an RSframe and may also be informed of the number of RS code parity databytes and the code size. Herein, the RS code is used to configure (orform) the RS frame. The RS frame decoder 1006 also refers to the RSframe-associated information in order to perform an inverse process ofthe RS frame encoder, which is included in the transmitting system,thereby correcting the errors within the RS frame. Thereafter, the RSframe decoder 1006 adds 1 MPEG synchronization data byte to theerror-correction mobile service data packet. In an earlier process, the1 MPEG synchronization data byte was removed from the mobile servicedata packet during the RS frame encoding process. Finally, the RS framedecoder 1006 outputs the processed mobile service data packet to thederandomizer 1007.

FIG. 64 illustrates, when the RS frame mode value is equal to ‘00’, anexemplary process of grouping several portion being transmitted to aparade, thereby forming an RS frame and an RS frame reliability map, andan exemplary process of performing a row de-permutation process in superframe units as an inverse process of the transmitting system, therebyre-distinguishing (or identifying) the row-de-permuted RS frame and RSframe reliability map. More specifically, the RS frame decoder 1006receives and groups a plurality of mobile service data bytes, so as toform an RS frame. According to the present invention, in transmittingsystem, the mobile service data correspond to data RS-encoded in RSframe units and also correspond to data row-permuted in super frameunits. At this point, the mobile service data may already be errorcorrection encoded (e.g., CRC-encoded). Alternatively, the errorcorrection encoding process may be omitted.

It is assumed that, in the transmitting system, an RS frame having thesize of (N+2)×(187+P) bytes is divided into M number of portions, andthat the M number of mobile service data portions are assigned andtransmitted to regions A/B/C/D in M number of data groups, respectively.In this case, in the receiving system, each mobile service data portionis grouped, as shown in FIG. 64( a), thereby forming an RS frame havingthe size of (N+2)×(187+P) bytes. At this point, when stuffing bytes (S)are added to at least one portion included in the corresponding RS frameand then transmitted, the stuffing bytes are removed, therebyconfiguring an RS frame and an RS frame reliability map. For example, asshown in FIG. 23, when S number of stuffing bytes are added to thecorresponding portion, the S number of stuffing bytes are removed,thereby configuring the RS frame and the RS frame reliability map.

Herein, when it is assumed that the block decoder 1005 outputs a softdecision value for the decoding result, the RS frame decoder 1006 maydecide the ‘0’ and ‘1’ of the corresponding bit by using the codes ofthe soft decision value. 8 bits that are each decided as described aboveare grouped to create 1 data byte. If the above-described process isperformed on all soft decision values of several portions (or datagroups) included in a parade, the RS frame having the size of(N+2)×(187+P) bytes may be configured.

Additionally, the present invention uses the soft decision value notonly to configure the RS frame but also to configure a reliability map.

Herein, the reliability map indicates the reliability of thecorresponding data byte, which is configured by grouping 8 bits, the 8bits being decided by the codes of the soft decision value.

For example, when the absolute value of the soft decision value exceedsa pre-determined threshold value, the value of the corresponding bit,which is decided by the code of the corresponding soft decision value,is determined to be reliable. Conversely, when the absolute value of thesoft decision value does not exceed the pre-determined threshold value,the value of the corresponding bit is determined to be unreliable.Thereafter, if even a single bit among the 8 bits, which are decided bythe codes of the soft decision value and group to configure one databyte, is determined to be unreliable, the corresponding data byte ismarked on the reliability map as an unreliable data byte.

Herein, determining the reliability of one data byte is only exemplary.More specifically, when a plurality of data bytes (e.g., at least 4 databytes) are determined to be unreliable, the corresponding data bytes mayalso be marked as unreliable data bytes within the reliability map.Conversely, when all of the data bits within the one data byte aredetermined to be reliable (i.e., when the absolute value of the softdecision values of all 8 bits included in the one data byte exceed thepredetermined threshold value), the corresponding data byte is marked tobe a reliable data byte on the reliability map. Similarly, when aplurality of data bytes (e.g., at least 4 data bytes) are determined tobe reliable, the corresponding data bytes may also be marked as reliabledata bytes within the reliability map. The numbers proposed in theabove-described example are merely exemplary and, therefore, do notlimit the scope or spirit of the present invention.

The process of configuring the RS frame and the process of configuringthe reliability map both using the soft decision value may be performedat the same time. Herein, the reliability information within thereliability map is in a one-to-one correspondence with each byte withinthe RS frame. For example, if a RS frame has the size of (N+2)×(187+P)bytes, the reliability map is also configured to have the size of(N+2)×(187+P) bytes. FIG. 64( a′) and FIG. 64( b′) respectivelyillustrate the process steps of configuring the reliability mapaccording to the present invention.

At this point, the RS frame of FIG. 64( b) and the RS frame reliabilitymap of FIG. 64( b′) are interleaved in super frame units (as shown inFIG. 21). Therefore, the RS frame and the RS frame reliability maps aregrouped to create a super frame and a super frame reliability map.Subsequently, as shown in FIG. 64( c) and FIG. 64( c′), a de-permutation(or deinterleaving) process is performed in super frame units on the RSframe and the RS frame reliability maps, as an inverse process of thetransmitting system. Then, when the de-permutation process is performedin super frame units, the processed data are divided into de-permuted(or deinterleaved) RS frames having the size of (N+2)×(187+P) bytes andde-permuted RS frame reliability maps having the size of (N+2)×(187+P)bytes, as shown in FIG. 64( d) and FIG. 64( d′). Subsequently, the RSframe reliability map is used on the divided RS frames so as to performerror correction.

FIG. 65 illustrates example of the error correction processed accordingto embodiments of the present invention. FIG. 65 illustrates an exampleof performing an error correction process when the transmitting systemhas performed both RS encoding and CRC encoding processes on the RSframe.

As shown in FIG. 65( a) and FIG. 65( a′), when the RS frame having thesize of (N+2)×(187+P) bytes and the RS frame reliability map having thesize of (N+2)×(187+P) bytes are created, a CRC syndrome checking processis performed on the created RS frame, thereby verifying whether anyerror has occurred in each row. Subsequently, as shown in FIG. 65( b), a2-byte checksum is removed to configure an RS frame having the size ofN×(187+P) bytes. Herein, the presence (or existence) of an error isindicated on an error flag corresponding to each row. Similarly, sincethe portion of the reliability map corresponding to the CRC checksum hashardly any applicability, this portion is removed so that only N×(187+P)number of the reliability information bytes remain, as shown in FIG. 65(b′).

After performing the CRC syndrome checking process, as described above,a RS decoding process is performed in a column direction. Herein, a RSerasure correction process may be performed in accordance with thenumber of CRC error flags. More specifically, as shown in FIG. 65( c),the CRC error flag corresponding to each row within the RS frame isverified. Thereafter, the RS frame decoder 1006 determines whether thenumber of rows having a CRC error occurring therein is equal to orsmaller than the maximum number of errors on which the RS erasurecorrection may be performed, when performing the RS decoding process ina column direction. The maximum number of errors corresponds to P numberof parity bytes inserted when performing the RS encoding process. In theembodiment of the present invention, it is assumed that 48 parity byteshave been added to each column (i.e., P=48).

If the number of rows having the CRC errors occurring therein is smallerthan or equal to the maximum number of errors (i.e., 48 errors accordingto this embodiment) that can be corrected by the RS erasure decodingprocess, a (235,187)-RS erasure decoding process is performed in acolumn direction on the RS frame having (187+P) number of N-byte rows(i.e., 235 N-byte rows), as shown in FIG. 65( d). Thereafter, as shownin FIG. 65( e), the 48-byte parity data that have been added at the endof each column are removed. Conversely, however, if the number of rowshaving the CRC errors occurring therein is greater than the maximumnumber of errors (i.e., 48 errors) that can be corrected by the RSerasure decoding process, the RS erasure decoding process cannot beperformed. In this case, the error may be corrected by performing ageneral RS decoding process. In addition, the reliability map, which hasbeen created based upon the soft decision value along with the RS frame,may be used to further enhance the error correction ability (orperformance) of the present invention.

More specifically, the RS frame decoder 1006 compares the absolute valueof the soft decision value of the block decoder 1005 with thepre-determined threshold value, so as to determine the reliability ofthe bit value decided by the code of the corresponding soft decisionvalue. Also, 8 bits, each being determined by the code of the softdecision value, are grouped to form one data byte. Accordingly, thereliability information on this one data byte is indicated on thereliability map. Therefore, as shown in FIG. 65( c), even though aparticular row is determined to have an error occurring therein basedupon a CRC syndrome checking process on the particular row, the presentinvention does not assume that all bytes included in the row have errorsoccurring therein. The present invention refers to the reliabilityinformation of the reliability map and sets only the bytes that havebeen determined to be unreliable as erroneous bytes. In other words,with disregard to whether or not a CRC error exists within thecorresponding row, only the bytes that are determined to be unreliablebased upon the reliability map are set as erasure points.

According to another method, when it is determined that CRC errors areincluded in the corresponding row, based upon the result of the CRCsyndrome checking result, only the bytes that are determined by thereliability map to be unreliable are set as errors. More specifically,only the bytes corresponding to the row that is determined to haveerrors included therein and being determined to be unreliable based uponthe reliability information, are set as the erasure points. Thereafter,if the number of error points for each column is smaller than or equalto the maximum number of errors (i.e., 48 errors) that can be correctedby the RS erasure decoding process, an RS erasure decoding process isperformed on the corresponding column. Conversely, if the number oferror points for each column is greater than the maximum number oferrors (i.e., 48 errors) that can be corrected by the RS erasuredecoding process, a general decoding process is performed on thecorresponding column.

More specifically, if the number of rows having CRC errors includedtherein is greater than the maximum number of errors (i.e., 48 errors)that can be corrected by the RS erasure decoding process, either an RSerasure decoding process or a general RS decoding process is performedon a column that is decided based upon the reliability information ofthe reliability map, in accordance with the number of erasure pointswithin the corresponding column. For example, it is assumed that thenumber of rows having CRC errors included therein within the RS frame isgreater than 48. And, it is also assumed that the number of erasurepoints decided based upon the reliability information of the reliabilitymap is indicated as 40 erasure points in the first column and as 50erasure points in the second column. In this case, a (235,187)-RSerasure decoding process is performed on the first column.Alternatively, a (235,187)-RS decoding process is performed on thesecond column. When error correction decoding is performed on all columndirections within the RS frame by using the above-described process, the48-byte parity data which were added at the end of each column areremoved, as shown in FIG. 65( e).

As described above, even though the total number of CRC errorscorresponding to each row within the RS frame is greater than themaximum number of errors that can be corrected by the RS erasuredecoding process, when the number of bytes determined to have a lowreliability level, based upon the reliability information on thereliability map within a particular column, while performing errorcorrection decoding on the particular column. Herein, the differencebetween the general RS decoding process and the RS erasure decodingprocess is the number of errors that can be corrected. Morespecifically, when performing the general RS decoding process, thenumber of errors corresponding to half of the number of parity bytes(i.e., (number of parity bytes)/2) that are inserted during the RSencoding process may be error corrected (e.g., 24 errors may becorrected). Alternatively, when performing the RS erasure decodingprocess, the number of errors corresponding to the number of paritybytes that are inserted during the RS encoding process may be errorcorrected (e.g., 48 errors may be corrected).

After performing the error correction decoding process, as describedabove, a RS frame configured of 187 N-byte rows (or packet) may beobtained as shown in FIG. 65( e). The RS frame having the size of N×187bytes is outputted by the order of N number of 187-byte units. At thispoint, 1 MPEG synchronization byte, which had been removed by thetransmitting system, is added to each 187-byte packet, as shown in FIG.65( f). Therefore, a 188-byte unit mobile service data packet isoutputted.

As described above, the RS frame decoded mobile service data isoutputted to the data derandomizer 1007. The data derandomizer 1007performs a derandomizing process, which corresponds to the inverseprocess of the randomizer included in the transmitting system, on thereceived mobile service data. Thereafter, the derandomized data areoutputted, thereby obtaining the mobile service data transmitted fromthe transmitting system. In the present invention, the RS frame decoder1006 may perform the data derandomizing function. An MPH frame decodermay be configured of M number of RS frame decoders provided in parallel,wherein the number of RS frame encoders is equal to the number ofparades (=M) within an MPH frame, a multiplexer for multiplexing eachportion and being provided to each input end of the M number of RS framedecoders, and a demultiplexer for demultiplexing each portion and beingprovided to each output end of the M number of RS frame decoders.

General Digital Broadcast Receiving System

FIG. 66 illustrates a block diagram showing a structure of a digitalbroadcast receiving system according to an embodiment of the presentinvention. Herein, the demodulating unit of FIG. 36 may be applied inthe digital broadcast receiving system. Referring to FIG. 66, thedigital broadcast receiving system includes a tuner 6001, a demodulatingunit 6002, a demultiplexer 6003, an audio decoder 6004, a video decoder6005, a native TV application manager 6006, a channel manager 6007, achannel map 6008, a first memory 6009, an SI and/or data decoder 6010, asecond memory 6011, a system manager 6012, a data broadcast applicationmanager 6013, a storage controller 6014, a third memory 6015, and a GPSmodule 6020. Herein, the first memory 6009 corresponds to a non-volatilerandom access memory (NVRAM) (or a flash memory). The third memory 6015corresponds to a large-scale storage device, such as a hard disk drive(HDD), a memory chip, and so on.

The tuner 6001 tunes a frequency of a specific channel through any oneof an antenna, cable, and satellite. Then, the tuner 6001 down-convertsthe tuned frequency to an intermediate frequency (IF), which is thenoutputted to the demodulating unit 6002. At this point, the tuner 6001is controlled by the channel manager 6007. Additionally, the result andstrength of the broadcast signal of the tuned channel are also reportedto the channel manager 6007. The data that are being received by thefrequency of the tuned specific channel include main service data,mobile service data, and table data for decoding the main service dataand mobile service data.

According to the embodiment of the present invention, audio data andvideo data for mobile broadcast programs may be applied as the mobileservice data. Such audio data and video data are compressed by varioustypes of encoders so as to be transmitted to a broadcasting station. Inthis case, the video decoder 6004 and the audio decoder 6005 will beprovided in the receiving system so as to correspond to each of theencoders used for the compression process. Thereafter, the decodingprocess will be performed by the video decoder 6004 and the audiodecoder 6005. Then, the processed video and audio data will be providedto the users. Examples of the encoding/decoding scheme for the audiodata may include AC 3, MPEG 2 AUDIO, MPEG 4 AUDIO, AAC, AAC+, HE AAC,AAC SBR, MPEG-Surround, and BSAC. And, examples of the encoding/decodingscheme for the video data may include MPEG 2 VIDEO, MPEG 4 VIDEO, H.264,SVC, and VC-1.

The audio decoder 6004 may extract audio data and supplementaryinformation from the audio stream demultiplexed by the demultiplexer6003. The audio decoder 6004 may inversely scale the extracted audiodata on the basis of the scale factor indicated by scale factor indexinformation included in the supplementary information and restore theaudio signal.

The audio decoder 6004 may inversely quantize the audio data, convertthe inversely quantized audio data into a time-domain signal for eachblock according to a block length indicated by an identifier included inthe supplementary information, and restore the audio signal.

The audio decoder 6004 may decode the audio data on the basis of Huffmantable information and region identifying information included in theaudio data and restore the audio signal.

The audio decoder 6004 may extract an audio parameter from the audiostream demultiplexed by the demultiplexer 6003. The audio decoder 6004may calculate subframe number information included in the audio streamon the basis of the extracted audio parameter and calculate startlocation information of the subframes on the basis of the calculatedsubframe number information. The audio decoder 6004 may decode the audiodata included in the subframes according to the audio parameter on thebasis of the calculated start location information.

The audio decoder 6004 may extract the audio data from the audio streamdemultiplexed by the demultiplexer 6003, inversely quantize the audiodata, and restore a stereo audio signal on the basis of mid data andside data included in the inversely quantized audio data.

The audio decoder 6004 may extract a parameter from the audio streamdemultiplexed by the demultiplexer 6003, decode the extracted parameter,and restore the audio signal in an extended region. The audio decoder6004 may decode the extracted parameter according to algebraic codeexcited linear prediction (ACELP) and transform coded excitation (TCX)and restore the audio signal in the extended region. If the parameterincludes spectral band replication (SBR), the audio decoder 6004 maydecode the parameter and restore the audio signal in a high frequencyregion of a mono channel.

Depending upon the embodiment of the present invention, examples of themobile service data may include data provided for data service, such asJava application data, HTML application data, XML data, and so on. Thedata provided for such data services may correspond either to a Javaclass file for the Java application, or to a directory file designatingpositions (or locations) of such files. Furthermore, such data may alsocorrespond to an audio file and/or a video file used in eachapplication. The data services may include weather forecast services,traffic information services, stock information services, servicesproviding information quiz programs providing audience participationservices, real time poll, user interactive education programs, gamingservices, services providing information on soap opera (or TV series)synopsis, characters, original sound track, filing sites, servicesproviding information on past sports matches, profiles andaccomplishments of sports players, product information and productordering services, services providing information on broadcast programsby media type, airing time, subject, and so on. The types of dataservices described above are only exemplary and are not limited only tothe examples given herein. Furthermore, depending upon the embodiment ofthe present invention, the mobile service data may correspond to metadata. For example, the meta data be written in XML format so as to betransmitted through a DSM-CC protocol.

The demodulating unit 6002 performs VSB-demodulation and channelequalization on the signal being outputted from the tuner 6001, therebyidentifying the main service data and the mobile service data.Thereafter, the identified main service data and mobile service data areoutputted in TS packet units. An example of the demodulating unit 6002is shown in FIG. 36 to FIG. 65. Therefore, the structure and operationof the demodulator will be described in detail in a later process.However, this is merely exemplary and the scope of the present inventionis not limited to the example set forth herein. In the embodiment givenas an example of the present invention, only the mobile service datapacket outputted from the demodulating unit 6002 is inputted to thedemultiplexer 6003. In this case, the main service data packet isinputted to another demultiplexer (not shown) that processes mainservice data packets. Herein, the storage controller 6014 is alsoconnected to the other demultiplexer in order to store the main servicedata after processing the main service data packets. The demultiplexerof the present invention may also be designed to process both mobileservice data packets and main service data packets in a singledemultiplexer.

The storage controller 6014 is interfaced with the demultipelxer so asto control instant recording, reserved (or pre-programmed) recording,time shift, and so on of the mobile service data and/or main servicedata. For example, when one of instant recording, reserved (orpre-programmed) recording, and time shift is set and programmed in thereceiving system (or receiver) shown in FIG. 66, the correspondingmobile service data and/or main service data that are inputted to thedemultiplexer are stored in the third memory 6015 in accordance with thecontrol of the storage controller 6014. The third memory 6015 may bedescribed as a temporary storage area and/or a permanent storage area.Herein, the temporary storage area is used for the time shiftingfunction, and the permanent storage area is used for a permanent storageof data according to the user's choice (or decision).

When the data stored in the third memory 6015 need to be reproduced (orplayed), the storage controller 6014 reads the corresponding data storedin the third memory 6015 and outputs the read data to the correspondingdemultiplexer (e.g., the mobile service data are outputted to thedemultiplexer 6003 shown in FIG. 66). At this point, according to theembodiment of the present invention, since the storage capacity of thethird memory 6015 is limited, the compression encoded mobile servicedata and/or main service data that are being inputted are directlystored in the third memory 6015 without any modification for theefficiency of the storage capacity. In this case, depending upon thereproduction (or reading) command, the data read from the third memory6015 pass trough the demultiplexer so as to be inputted to thecorresponding decoder, thereby being restored to the initial state.

The storage controller 6014 may control the reproduction (or play),fast-forward, rewind, slow motion, instant replay functions of the datathat are already stored in the third memory 6015 or presently beingbuffered. Herein, the instant replay function corresponds to repeatedlyviewing scenes that the viewer (or user) wishes to view once again. Theinstant replay function may be performed on stored data and also on datathat are currently being received in real time by associating theinstant replay function with the time shift function. If the data beinginputted correspond to the analog format, for example, if thetransmission mode is NTSC, PAL, and so on, the storage controller 6014compression encodes the inputted data and stored the compression-encodeddata to the third memory 6015. In order to do so, the storage controller6014 may include an encoder, wherein the encoder may be embodied as oneof software, middleware, and hardware. Herein, an MPEG encoder may beused as the encoder according to an embodiment of the present invention.The encoder may also be provided outside of the storage controller 6014.

Meanwhile, in order to prevent illegal duplication (or copies) of theinput data being stored in the third memory 6015, the storage controller6014 scrambles (or encrypts) the input data and stores the scrambled (orencrypted) data in the third memory 6015. Accordingly, the storagecontroller 6014 may include a scramble algorithm (or encryptionalgorithm) for scrambling the data stored in the third memory 6015 and adescramble algorithm (or decryption algorithm) for descrambling (ordecrypting) the data read from the third memory 6015. The scramblingmethod may include using an arbitrary key (e.g., control word) to modifya desired set of data, and also a method of mixing signals.

Meanwhile, the demultiplexer 6003 receives the real-time data outputtedfrom the demodulating unit 6002 or the data read from the third memory6015 and demultiplexes the received data. In the example given in thepresent invention, the demultiplexer 6003 performs demultiplexing on themobile service data packet. Therefore, in the present invention, thereceiving and processing of the mobile service data will be described indetail. However, depending upon the many embodiments of the presentinvention, not only the mobile service data but also the main servicedata may be processed by the demultiplexer 6003, the audio decoder 6004,the video decoder 6005, the native TV application manager 6006, thechannel manager 6007, the channel map 6008, the first memory 6009, theSI and/or data decoder 6010, the second memory 6011, a system manager6012, the data broadcast application manager 6013, the storagecontroller 6014, the third memory 6015, and the GPS module 6020.Thereafter, the processed data may be used to provide diverse servicesto the users.

The demultiplexer 6003 demultiplexes mobile service data and systeminformation (SI) tables from the mobile service data packet inputted inaccordance with the control of the SI and/or data decoder 6010.Thereafter, the demultiplexed mobile service data and SI tables areoutputted to the SI and/or data decoder 6010 in a section format. Inthis case, it is preferable that data for the data service are used asthe mobile service data that are inputted to the SI and/or data decoder6010. In order to extract the mobile service data from the channelthrough which mobile service data are transmitted and to decode theextracted mobile service data, system information is required. Suchsystem information may also be referred to as service information. Thesystem information may include channel information, event information,etc. In the embodiment of the present invention, the PSI/PSIP tables areapplied as the system information. However, the present invention is notlimited to the example set forth herein. More specifically, regardlessof the name, any protocol transmitting system information in a tableformat may be applied in the present invention.

The PSI table is an MPEG-2 system standard defined for identifying thechannels and the programs. The PSIP table is an advanced televisionsystems committee (ATSC) standard that can identify the channels and theprograms. The PSI table may include a program association table (PAT), aconditional access table (CAT), a program map table (PMT), and a networkinformation table (NIT). Herein, the PAT corresponds to specialinformation that is transmitted by a data packet having a PID of ‘0’.The PAT transmits PID information of the PMT and PID information of theNIT corresponding to each program. The CAT transmits information on apaid broadcast system used by the transmitting system. The PMT transmitsPID information of a transport stream (TS) packet, in which programidentification numbers and individual bit sequences of video and audiodata configuring the corresponding program are transmitted, and the PIDinformation, in which PCR is transmitted. The NIT transmits informationof the actual transmission network.

The PSIP table may include a virtual channel table (VCT), a system timetable (STT), a rating region table (RRT), an extended text table (ETT),a direct channel change table (DCCT), an event information table (EIT),and a master guide table (MGT). The VCT transmits information on virtualchannels, such as channel information for selecting channels andinformation such as packet identification (PID) numbers for receivingthe audio and/or video data. More specifically, when the VCT is parsed,the PID of the audio/video data of the broadcast program may be known.Herein, the corresponding audio/video data are transmitted within thechannel along with the channel name and the channel number.

FIG. 67 illustrates a VCT syntax according to an embodiment of thepresent invention. The VCT syntax of FIG. 67 is configured by includingat least one of a table_id field, a section_syntax_indicator field, aprivate_indicator field, a section_length field, a transport_stream idfield, a version_number field, a current_next_indicator field, asection_number field, a last_section_number field, a protocol_versionfield, and a num_channels_in_section field.

The VCT syntax further includes a first ‘for’ loop repetition statementthat is repeated as much as the num_channels_in_section field value. Thefirst repetition statement may include at least one of a short_namefield, a major_channel_number field, a minor_channel_number field, amodulation_mode field, a carrier_frequency field, a channel_TSID field,a program_number field, an ETM_location field, an access_controlledfield, a hidden field, a service_type field, a source_id field, adescriptor_length field, and a second ‘for’ loop statement that isrepeated as much as the number of descriptors included in the firstrepetition statement. Herein, the second repetition statement will bereferred to as a first descriptor loop for simplicity. The descriptordescriptors( ) included in the first descriptor loop is separatelyapplied to each virtual channel.

Furthermore, the VCT syntax may further include anadditional_descriptor_length field, and a third ‘for’ loop statementthat is repeated as much as the number of descriptors additionally addedto the VCT. For simplicity of the description of the present invention,the third repetition statement will be referred to as a seconddescriptor loop. The descriptor additional_descriptors( ) included inthe second descriptor loop is commonly applied to all virtual channelsdescribed in the VCT.

As described above, referring to FIG. 67, the table_id field indicates aunique identifier (or identification) (ID) that can identify theinformation being transmitted to the table as the VCT. Morespecifically, the table_id field indicates a value informing that thetable corresponding to this section is a VCT. For example, a 0xC8 valuemay be given to the table_id field.

The version_number field indicates the version number of the VCT. Thesection_number field indicates the number of this section. Thelast_section_number field indicates the number of the last section of acomplete VCT. And, the num_channel_in_section field designates thenumber of the overall virtual channel existing within the VCT section.Furthermore, in the first ‘for’ loop repetition statement, theshort_name field indicates the name of a virtual channel. Themajor_channel_number field indicates a ‘major’ channel number associatedwith the virtual channel defined within the first repetition statement,and the minor_channel_number field indicates a ‘minor’ channel number.More specifically, each of the channel numbers should be connected tothe major and minor channel numbers, and the major and minor channelnumbers are used as user reference numbers for the corresponding virtualchannel.

The program_number field is shown for connecting the virtual channelhaving an MPEG-2 program association table (PAT) and program map table(PMT) defined therein, and the program_number field matches the programnumber within the PAT/PMT. Herein, the PAT describes the elements of aprogram corresponding to each program number, and the PAT indicates thePID of a transport packet transmitting the PMT. The PMT describedsubordinate information, and a PID list of the transport packet throughwhich a program identification number and a separate bit sequence, suchas video and/or audio data configuring the program, are beingtransmitted.

FIG. 68 illustrates a service_type field according to an embodiment ofthe present invention. The service_type field indicates the service_typeprovided in a corresponding virtual channel. Referring to FIG. 68, it isprovided that the service_type field should only indicate an analogtelevision, a digital television, digital audio data, and digital videodata. Also, according to the embodiment of the present invention, it maybe provided that a mobile broadcast program should be designated to theservice_type field. The service_type field, which is parsed by the SIand/or data decoder 6010 may be provided to a receiving system, as shownin FIG. 66, and used accordingly. According to other embodiments of thepresent invention, the parsed service_type field may also be provided toeach of the audio decoder 6004 and video decoder 6005, so as to be usedin the decoding process.

The source_id field indicates a program source connected to thecorresponding virtual channel. Herein, a source refers to a specificsource, such as an image, a text, video data, or sound. The source_idfield value has a unique value within the transport stream transmittingthe VCT. Meanwhile, a service location descriptor may be included in adescriptor loop (i.e., descriptor{ }) within a next ‘for’ looprepetition statement. The service location descriptor may include astream type, PID, and language code for each elementary stream.

FIG. 69 illustrates a service location descriptor according to anembodiment of the present invention. As shown in FIG. 69, the servicelocation descriptor may include a descriptor_tag field, adescriptor_length field, and a PCR_PID field. Herein, the PCR_PID fieldindicates the PID of a transport stream packet within a programspecified by a program number field, wherein the transport stream packetincludes a valid PCR field. Meanwhile, the service location descriptorincludes a number_elements field so as to indicate a number of PIDs usedin the corresponding program. The number of repetition of a next ‘for’descriptor loop repetition statement can be decided, depending upon thevalue of the number_elements field. Referring to FIG. 69, the ‘for’ looprepetition statement includes a stream_type field, an elementary_PIDfield, and an ISO_(—)639_language_code field. Herein, the stream_typefield indicates the stream type of the corresponding elementary stream(i.e., video/audio data). The elementary_PID field indicates the PID ofthe corresponding elementary stream. The ISO_(—)639_language_code fieldindicates a language code of the corresponding elementary stream.

FIG. 70 illustrates examples that may be assigned to the stream_typefield according to the present invention. As shown in FIG. 70, ISO/IEC11172 Video, ITU-T Rec. H.262|ISO/IEC 13818-2 Video or ISO/IEC 11172-2constrained parameter video stream, ISO/IEC 11172 Audio, ISO/IEC 13818-3Audio, ITU-T Rec. H.222.01 ISO/IEC 13818-1 private_sections, ITU-T Rec.H.222.01 ISO/IEC 13818-1 PES packets containing private data, ISO/IEC13522 MHEG, ITU-T Rec. H.222.0|ISO/IEC 13818-1 Annex A DSM CC, ITU-TRec. H.222.1, ISO/IEC 13818-6 type A, ISO/IEC 13818-6 type B, ISO/IEC13818-6 type C, ISO/IEC 13818-6 type D, ISO/IEC 13818-1 auxiliary, andso on may be applied as the stream type. Meanwhile, according to theembodiment of the present invention, MPH video stream: Non-hierarchicalmode, MPH audio stream: Non-hierarchical mode, MPH Non-A/V stream:Non-hierarchical mode, MPH High Priority video stream: Hierarchicalmode, MPH High Priority audio stream: Hierarchical mode, MPH LowPriority video stream: Hierarchical mode, MPH Low priority audio stream:Hierarchical mode, and so on may further be applied as the stream type.

As described above, “MPH” corresponds to the initials of “mobile”,“pedestrian”, and “handheld” and represents the opposite concept of afixed-type system. Therefore, the MPH video stream: Non-hierarchicalmode, the MPH audio stream: Non-hierarchical mode, the MPH Non-A/Vstream: Non-hierarchical mode, the MPH High Priority video stream:Hierarchical mode, the MPH High Priority audio stream: Hierarchicalmode, the MPH Low Priority video stream: Hierarchical mode, and the MPHLow priority audio stream: Hierarchical mode correspond to stream typesthat are applied when mobile broadcast programs are being transmittedand received. Also the Hierarchical mode and the Non-hierarchical modeeach correspond to values that are used in stream types having differentpriority levels. Herein, the priority level is determined based upon ahierarchical structure applied in any one of the encoding or decodingmethod.

Therefore, when a hierarchical structure-type codec is used, a fieldvalue including the hierarchical mode and the non-hierarchical mode isrespectively designated so as to identify each stream. Such stream typeinformation is parsed by the SI and/or data decoder 6010, so as to beprovided to the video and audio decoders 6004 and 6005. Thereafter, eachof the video and audio decoders 6004 and 6005 uses the parsed streamtype information in order to perform the decoding process. Other streamtypes that may be applied in the present invention may include MPEG 4AUDIO, AC 3, AAC, AAC+, BSAC, HE AAC, AAC SBR, and MPEG-S for the audiodata, and may also include MPEG 2 VIDEO, MPEG 4 VIDEO, H.264, SVC, andVC-1 for the video data.

Furthermore, referring to FIG. 70, in fields using the hierarchical modeand the non-hierarchical mode, such as the MPH video stream:Non-hierarchical mode and the MPH audio stream: Non-hierarchical mode,examples of using the MPEG 4 AUDIO, AC 3, AAC, AAC+, BSAC, HE AAC, AACSBR, and MPEG-S for the audio data, and the MPEG 2 VIDEO, MPEG 4 VIDEO,H.264, SVC, and VC-1 for the video data may also be respectively used asreplacements for each of the audio stream and the video stream may beconsidered as other embodiments of the present invention and may,therefore, be included in the scope of the present invention. Meanwhile,the stream_type field may be provided as one of the fields within thePMT. And, in this case, it is apparent that such stream_type fieldincludes the above-described syntax. The STT transmits information onthe current data and timing information. The RRT transmits informationon region and consultation organs for program ratings. The ETT transmitsadditional description of a specific channel and broadcast program. TheEIT transmits information on virtual channel events (e.g., programtitle, program start time, etc.).

FIG. 71 illustrates a bit stream syntax for an event information table(EIT) according to the present invention. In this embodiment, the EITshown in FIG. 71 corresponds to a PSIP table including information on atitle, start time, duration, and so on of an event in a virtual channel.Referring to FIG. 71, the EIT is configured of a plurality of fieldsincluding a table_id field, a section_syntax_indicator field, aprivate_indicator field, a source_ID, a version_numbers_in_sectionfield, a current_next_indicator field, and a num_event field. Morespecifically, the table_id field is an 8-bit field having the value of‘oxCB’, which indicates that the corresponding section is included inthe EIT. The section_syntax_indicator field is a 1-bit field having thevalue of ‘1’. This indicates that the corresponding section passesthrough the section_length field and is in accordance with a genericsection syntax. The private_indicator field corresponds to a 1-bit fieldhaving the value of ‘1’.

Also, the source_ID corresponds to an ID identifying a virtual channelthat carries an event shown in the above-described table. The versionnumbers in section field indicates the version of an element included inthe event information table. In the present invention, with respect tothe previous version number, an event change information included in theevent information table, wherein the event change information has a newversion number is recognized as the latest change in information. Thecurrent_next_indicator field indicates whether the event informationincluded in the corresponding EIT is a current information or a nextinformation. And, finally, the num_event field represents the number ofevents included in the channel having a source ID. More specifically, anevent loop shown below is repeated as many times as the number ofevents.

The above-described EIT field is commonly applied to at least one ormore events included in one EIT syntax. A loop statement, which isincluded as “for (j=0;j<num_event_in_section;j++){ }”, describes thecharacteristics of each event. The following fields represent detailedinformation of each individual event. Therefore, the following fieldsare individually applied to each corresponding event described by theEIT syntax. An event_ID included in an event loop is an identifier foridentifying each individual event. The number of the event IDcorresponds to a portion of the identifier for even extended textmessage (i.e., ETM_ID). A start_time field indicates the starting timeof an event. Therefore, the start_time field collects the starting timeinformation of a program provided from an electronic programinformation. A length_in_seconds field indicates the duration of anevent. Therefore, the length_in_seconds field collects the ending timeinformation of a program provided from an electronic programinformation. More specifically, the ending time information is collectedby adding the start_time field value and the length_in_secodns fieldvalue. A title_text( ) field may be used to indicate the tile of abroadcast program.

Meanwhile, the descriptor applied to each event may be included in theEIT. Herein, a descriptors_length field indicates the length of adescriptor. Also, a descriptor loop (i.e., descriptor{ }) included in a‘for’ loop repetition statement includes at least one of an AC-3 audiodescriptor, an MPEG 2 audio descriptor, an MPEG 4 audio descriptor, anAAC descriptor, an AAC+descriptor, an HE AAC descriptor, an AAC SBRdescriptor, an MPEG surround descriptor, a BSAC descriptor, an MPEG 2video descriptor, an MPEG 4 video descriptor, an H.264 descriptor, anSVC descriptor, and a VC-1 descriptor. Herein, each descriptor describesinformation on audio/video codec applied to each event. Such codecinformation may be provided to the audio/video decoder 6004 and 6005 andused in the decoding process.

Finally, the DCCT/DCCSCT transmits information associated with automatic(or direct) channel change. And, the MGT transmits the versions and PIDinformation of the above-mentioned tables included in the PSIP. Each ofthe above-described tables included in the PSI/PSIP is configured of abasic unit referred to as a “section”, and a combination of one or moresections forms a table. For example, the VCT may be divided into 256sections. Herein, one section may include a plurality of virtual channelinformation. However, a single set of virtual channel information is notdivided into two or more sections. At this point, the receiving systemmay parse and decode the data for the data service that are transmittingby using only the tables included in the PSI, or only the tablesincluded in the PSIP, or a combination of tables included in both thePSI and the PSIP. In order to parse and decode the mobile service data,at least one of the PAT and PMT included in the PSI, and the VCTincluded in the PSIP is required. For example, the PAT may include thesystem information for transmitting the mobile service data, and the PIDof the PMT corresponding to the mobile service data (or program number).The PMT may include the PID of the TS packet used for transmitting themobile service data. The VCT may include information on the virtualchannel for transmitting the mobile service data, and the PID of the TSpacket for transmitting the mobile service data.

Meanwhile, depending upon the embodiment of the present invention, aDVB-SI may be applied instead of the PSIP. The DVB-SI may include anetwork information table (NIT), a service description table (SDT), anevent information table (EIT), and a time and data table (TDT). TheDVB-SI may be used in combination with the above-described PSI. Herein,the NIT divides the services corresponding to particular networkproviders by specific groups. The NIT includes all tuning informationthat are used during the IRD set-up. The NIT may be used for informingor notifying any change in the tuning information. The SDT includes theservice name and different parameters associated with each servicecorresponding to a particular MPEG multiplex. The EIT is used fortransmitting information associated with all events occurring in theMPEG multiplex. The EIT includes information on the current transmissionand also includes information selectively containing differenttransmission streams that may be received by the IRD. And, the TDT isused for updating the clock included in the IRD.

Furthermore, three selective SI tables (i.e., a bouquet associate table(BAT), a running status table (RST), and a stuffing table (ST)) may alsobe included. More specifically, the bouquet associate table (BAT)provides a service grouping method enabling the IRD to provide servicesto the viewers. Each specific service may belong to at least one‘bouquet’ unit. A running status table (RST) section is used forpromptly and instantly updating at least one event execution status. Theexecution status section is transmitted only once at the changing pointof the event status. Other SI tables are generally transmitted severaltimes. The stuffing table (ST) may be used for replacing or discarding asubsidiary table or the entire SI tables.

In the present invention, when the mobile service data correspond toaudio data and video data, it is preferable that the mobile service dataincluded (or loaded) in a payload within a TS packet correspond to PEStype mobile service data. According to another embodiment of the presentinvention, when the mobile service data correspond to the data for thedata service (or data service data), the mobile service data included inthe payload within the TS packet consist of a digital storagemedia-command and control (DSM-CC) section format. However, the TSpacket including the data service data may correspond either to apacketized elementary stream (PES) type or to a section type. Morespecifically, either the PES type data service data configure the TSpacket, or the section type data service data configure the TS packet.The TS packet configured of the section type data will be given as theexample of the present invention. At this point, the data service dataare includes in the digital storage media-command and control (DSM-CC)section. Herein, the DSM-CC section is then configured of a 188-byteunit TS packet.

Furthermore, the packet identification of the TS packet configuring theDSM-CC section is included in a data service table (DST). Whentransmitting the DST, ‘0x95’ is assigned as the value of a stream-typefield included in the service location descriptor of the PMT or the VCT.More specifically, when the PMT or VCT stream_type field value is‘0x95’, the receiving system may acknowledge the reception of the databroadcast program including mobile service data. At this point, themobile service data may be transmitted by a data/object carousel method.The data/object carousel method corresponds to repeatedly transmittingidentical data on a regular basis.

At this point, according to the control of the SI and/or data decoder6010, the demultiplexer 6003 performs section filtering, therebydiscarding repetitive sections and outputting only the non-repetitivesections to the SI and/or data decoder 6010. The demultiplexer 6003 mayalso output only the sections configuring desired tables (e.g., VCT orEIT) to the SI and/or data decoder 6010 by section filtering. Herein,the VCT or EIT may include a specific descriptor for the mobile servicedata. However, the present invention does not exclude the possibilitiesof the mobile service data being included in other tables, such as thePMT. The section filtering method may include a method of verifying thePID of a table defined by the MGT, such as the VCT, prior to performingthe section filtering process. Alternatively, the section filteringmethod may also include a method of directly performing the sectionfiltering process without verifying the MGT, when the VCT includes afixed PID (i.e., a base PID). At this point, the demultiplexer 6003performs the section filtering process by referring to a table_id field,a version_number field, a section_number field, etc.

As described above, the method of defining the PID of the VCT broadlyincludes two different methods. Herein, the PID of the VCT is a packetidentifier required for identifying the VCT from other tables. The firstmethod consists of setting the PID of the VCT so that it is dependent tothe MGT. In this case, the receiving system cannot directly verify theVCT among the many PSI and/or PSIP tables. Instead, the receiving systemmust check the PID defined in the MGT in order to read the VCT. Herein,the MGT defines the PID, size, version number, and so on, of diversetables. The second method consists of setting the PID of the VCT so thatthe PID is given a base PID value (or a fixed PID value), thereby beingindependent from the MGT. In this case, unlike in the first method, theVCT according to the present invention may be identified without havingto verify every single PID included in the MGT. Evidently, an agreementon the base PID must be previously made between the transmitting systemand the receiving system.

Meanwhile, in the embodiment of the present invention, the demultiplexer6003 may output only an application information table (AIT) to the SIand/or data decoder 6010 by section filtering. The AIT includesinformation on an application being operated in the receiver for thedata service. The AIT may also be referred to as an XAIT, and an AMT.Therefore, any table including application information may correspond tothe following description. When the AIT is transmitted, a value of‘0x05’ may be assigned to a stream_type field of the PMT. The AIT mayinclude application information, such as application name, applicationversion, application priority, application ID, application status (i.e.,auto-start, user-specific settings, kill, etc.), application type (i.e.,Java or HTML), position (or location) of stream including applicationclass and data files, application platform directory, and location ofapplication icon.

In the method for detecting application information for the data serviceby using the AIT, component_tag, original_network_id,transport_stream_id, and service_id fields may be used for detecting theapplication information. The component_tag field designates anelementary stream carrying a DSI of a corresponding object carousel. Theoriginal_network_id field indicates a DVB-SI original_network_id of theTS providing transport connection. The transport_stream_id fieldindicates the MPEG TS of the TS providing transport connection, and theservice_id field indicates the DVB-SI of the service providing transportconnection. Information on a specific channel may be obtained by usingthe original_network_id field, the transport_stream_id field, and theservice_id field. The data service data, such as the application data,detected by using the above-described method may be stored in the secondmemory 6011 by the SI and/or data decoder 6010.

The SI and/or data decoder 6010 parses the DSM-CC section configuringthe demultiplexed mobile service data. Then, the mobile service datacorresponding to the parsed result are stored as a database in thesecond memory 6011. The SI and/or data decoder 6010 groups a pluralityof sections having the same table identification (table_id) so as toconfigure a table, which is then parsed. Thereafter, the parsed resultis stored as a database in the second memory 6011. At this point, byparsing data and/or sections, the SI and/or data decoder 6010 reads allof the remaining actual section data that are not section-filtered bythe demultiplexer 6003. Then, the SI and/or data decoder 6010 stores theread data to the second memory 6011. The second memory 6011 correspondsto a table and data/object carousel database storing system informationparsed from tables and mobile service data parsed from the DSM-CCsection. Herein, a table_id field, a section_number field, and alast_section_number field included in the table may be used to indicatewhether the corresponding table is configured of a single section or aplurality of sections. For example, TS packets having the PID of the VCTare grouped to form a section, and sections having table identifiersallocated to the VCT are grouped to form the VCT. When the VCT isparsed, information on the virtual channel to which mobile service dataare transmitted may be obtained.

Also, according to the present invention, the SI and/or data decoder6010 parses the SLD of the VCT, thereby transmitting the stream typeinformation of the corresponding elementary stream to the audio decoder6004 or the video decoder 6005. In this case, the corresponding audiodecoder 6004 or video decoder 6005 uses the transmitted stream typeinformation so as to perform the audio or video decoding process.Furthermore, according to the present invention, the SI and/or datadecoder 6010 parses an AC-3 audio descriptor, an MPEG 2 audiodescriptor, an MPEG 4 audio descriptor, an AAC descriptor, anAAC+descriptor, an HE AAC descriptor, an AAC SBR descriptor, an MPEGsurround descriptor, a BSAC descriptor, an MPEG 2 video descriptor, anMPEG 4 video descriptor, an H.264 descriptor, an SVC descriptor, a VC-1descriptor, and so on, of the EIT, thereby transmitting the audio orvideo codec information of the corresponding event to the audio decoder6004 or video decoder 6005. In this case, the corresponding audiodecoder 6004 or video decoder 6005 uses the transmitted audio or videocodec information in order to perform an audio or video decodingprocess.

The obtained application identification information, service componentidentification information, and service information corresponding to thedata service may either be stored in the second memory 6011 or beoutputted to the data broadcasting application manager 6013. Inaddition, reference may be made to the application identificationinformation, service component identification information, and serviceinformation in order to decode the data service data. Alternatively,such information may also prepare the operation of the applicationprogram for the data service. Furthermore, the SI and/or data decoder6010 controls the demultiplexing of the system information table, whichcorresponds to the information table associated with the channel andevents. Thereafter, an A/V PID list may be transmitted to the channelmanager 6007.

The channel manager 6007 may refer to the channel map 6008 in order totransmit a request for receiving system-related information data to theSI and/or data decoder 6010, thereby receiving the corresponding result.In addition, the channel manager 6007 may also control the channeltuning of the tuner 6001. Furthermore, the channel manager 6007 maydirectly control the demultiplexer 6003, so as to set up the A/V PID,thereby controlling the audio decoder 6004 and the video decoder 6005.

The audio decoder 6004 and the video decoder 6005 may respectivelydecode and output the audio data and video data demultiplexed from themain service data packet. Alternatively, the audio decoder 6004 and thevideo decoder 6005 may respectively decode and output the audio data andvideo data demultiplexed from the mobile service data packet. Meanwhile,when the mobile service data include data service data, and also audiodata and video data, it is apparent that the audio data and video datademultiplexed by the demultiplexer 6003 are respectively decoded by theaudio decoder 6004 and the video decoder 6005. For example, anaudio-coding (AC)-3 decoding algorithm, an MPEG-2 audio decodingalgorithm, an MPEG-4 audio decoding algorithm, an AAC decodingalgorithm, an AAC+decoding algorithm, an HE AAC decoding algorithm, anAAC SBR decoding algorithm, an MPEG surround decoding algorithm, and aBSAC decoding algorithm may be applied to the audio decoder 6004. Also,an MPEG-2 video decoding algorithm, an MPEG-4 video decoding algorithm,an H.264 decoding algorithm, an SVC decoding algorithm, and a VC-1decoding algorithm may be applied to the video decoder 6005.Accordingly, the decoding process may be performed.

Meanwhile, the native TV application manager 6006 operates a nativeapplication program stored in the first memory 6009, thereby performinggeneral functions such as channel change. The native application programrefers to software stored in the receiving system upon shipping of theproduct. More specifically, when a user request (or command) istransmitted to the receiving system through a user interface (UI), thenative TV application manger 6006 displays the user request on a screenthrough a graphic user interface (GUI), thereby responding to the user'srequest. The user interface receives the user request through an inputdevice, such as a remote controller, a key pad, a jog controller, an atouch-screen provided on the screen, and then outputs the received userrequest to the native TV application manager 6006 and the databroadcasting application manager 6013. Furthermore, the native TVapplication manager 6006 controls the channel manager 6007, therebycontrolling channel-associated operations, such as the management of thechannel map 6008, and controlling the SI and/or data decoder 6010. Thenative TV application manager 6006 also controls the GUI of the overallreceiving system, thereby storing the user request and status of thereceiving system in the first memory 6009 and restoring the storedinformation.

The channel manager 6007 controls the tuner 6001 and the SI and/or datadecoder 6010, so as to managing the channel map 6008 so that it canrespond to the channel request made by the user. More specifically,channel manager 6007 sends a request to the SI and/or data decoder 6010so that the tables associated with the channels that are to be tuned areparsed. The results of the parsed tables are reported to the channelmanager 6007 by the SI and/or data decoder 6010. Thereafter, based onthe parsed results, the channel manager 6007 updates the channel map6008 and sets up a PID in the demultiplexer 6003 for demultiplexing thetables associated with the data service data from the mobile servicedata.

The system manager 6012 controls the booting of the receiving system byturning the power on or off. Then, the system manager 6012 stores ROMimages (including downloaded software images) in the first memory 6009.More specifically, the first memory 6009 stores management programs suchas operating system (OS) programs required for managing the receivingsystem and also application program executing data service functions.The application program is a program processing the data service datastored in the second memory 6011 so as to provide the user with the dataservice. If the data service data are stored in the second memory 6011,the corresponding data service data are processed by the above-describedapplication program or by other application programs, thereby beingprovided to the user. The management program and application programstored in the first memory 6009 may be updated or corrected to a newlydownloaded program. Furthermore, the storage of the stored managementprogram and application program is maintained without being deleted evenif the power of the system is shut down. Therefore, when the power issupplied, the programs may be executed without having to be newlydownloaded once again.

The application program for providing data service according to thepresent invention may either be initially stored in the first memory6009 upon the shipping of the receiving system, or be stored in thefirst memory 6009 after being downloaded. The application program forthe data service (i.e., the data service providing application program)stored in the first memory 6009 may also be deleted, updated, andcorrected. Furthermore, the data service providing application programmay be downloaded and executed along with the data service data eachtime the data service data are being received.

When a data service request is transmitted through the user interface,the data broadcasting application manager 6013 operates thecorresponding application program stored in the first memory 6009 so asto process the requested data, thereby providing the user with therequested data service. And, in order to provide such data service, thedata broadcasting application manager 6013 supports the graphic userinterface (GUI). Herein, the data service may be provided in the form oftext (or short message service (SMS)), voice message, still image, andmoving image. The data broadcasting application manager 6013 may beprovided with a platform for executing the application program stored inthe first memory 6009. The platform may be, for example, a Java virtualmachine for executing the Java program. Hereinafter, an example of thedata broadcasting application manager 6013 executing the data serviceproviding application program stored in the first memory 6009, so as toprocess the data service data stored in the second memory 6011, therebyproviding the user with the corresponding data service will now bedescribed in detail.

Assuming that the data service corresponds to a traffic informationservice, the data service according to the present invention is providedto the user of a receiver that is not equipped with an electronic mapand/or a GPS system in the form of at least one of a text (or shortmessage service (SMS)), a voice message, a graphic message, a stillimage, and a moving image. In this case, when a GPS module 6020 ismounted on the receiving system, as shown in FIG. 66, the GPS module6020 receives satellite signals transmitted from a plurality of lowearth orbit satellites and extracts the current position (or location)information (e.g., longitude, latitude, altitude), thereby outputtingthe extracted information to the data broadcasting application manager6013.

At this point, it is assumed that the electronic map includinginformation on each link and nod and other diverse graphic informationare stored in one of the second memory 6011, the first memory 6009, andanother memory that is not shown. More specifically, according to therequest made by the data broadcasting application manager 6013, the dataservice data stored in the second memory 6011 are read and inputted tothe data broadcasting application manager 6013. The data broadcastingapplication manager 6013 translates (or deciphers) the data service dataread from the second memory 6011, thereby extracting the necessaryinformation according to the contents of the message and/or a controlsignal. In other words, the data broadcasting application manager 6013uses the current position information and the graphic information, sothat the current position information can be processed and provided tothe user in a graphic format.

FIG. 72 illustrates a block diagram showing the structure of a digitalbroadcast (or television) receiving system according to anotherembodiment of the present invention. Referring to FIG. 72, the digitalbroadcast receiving system includes a tuner 7001, a demodulating unit7002, a demultiplexer 7003, a first descrambler 7004, an audio decoder7005, a video decoder 7006, a second descrambler 7007, an authenticationunit 7008, a native TV application manager 7009, a channel manager 7010,a channel map 7011, a first memory 7012, a data decoder 7013, a secondmemory 7014, a system manager 7015, a data broadcasting applicationmanager 7016, a storage controller 7017, a third memory 7018, atelecommunication module 7019, and a GPS module 7020. Herein, the thirdmemory 7018 is a mass storage device, such as a hard disk drive (HDD) ora memory chip. Also, during the description of the digital broadcast (ortelevision or DTV) receiving system shown in FIG. 72, the componentsthat are identical to those of the digital broadcast receiving system ofFIG. 66 will be omitted for simplicity.

As described above, in order to provide services for preventing illegalduplication (or copies) or illegal viewing of the enhanced data and/ormain data that are transmitted by using a broadcast network, and toprovide paid broadcast services, the transmitting system may generallyscramble and transmit the broadcast contents. Therefore, the receivingsystem needs to descramble the scrambled broadcast contents in order toprovide the user with the proper broadcast contents. Furthermore, thereceiving system may generally be processed with an authenticationprocess with an authentication means before the descrambling process.Hereinafter, the receiving system including an authentication means anda descrambling means according to an embodiment of the present inventionwill now be described in detail.

According to the present invention, the receiving system may be providedwith a descrambling means receiving scrambled broadcasting contents andan authentication means authenticating (or verifying) whether thereceiving system is entitled to receive the descrambled contents.Hereinafter, the descrambling means will be referred to as first andsecond descramblers 7004 and 7007, and the authentication means will bereferred to as an authentication unit 7008. Such naming of thecorresponding components is merely exemplary and is not limited to theterms suggested in the description of the present invention. Forexample, the units may also be referred to as a decryptor. Although FIG.72 illustrates an example of the descramblers 7004 and 7007 and theauthentication unit 7008 being provided inside the receiving system,each of the descramblers 7004 and 7007 and the authentication unit 7008may also be separately provided in an internal or external module.Herein, the module may include a slot type, such as a SD or CF memory, amemory stick type, a USB type, and so on, and may be detachably fixed tothe receiving system.

As described above, when the authentication process is performedsuccessfully by the authentication unit 7008, the scrambled broadcastingcontents are descrambled by the descramblers 7004 and 7007, therebybeing provided to the user. At this point, a variety of theauthentication method and descrambling method may be used herein.However, an agreement on each corresponding method should be madebetween the receiving system and the transmitting system. Hereinafter,the authentication and descrambling methods will now be described, andthe description of identical components or process steps will be omittedfor simplicity.

The receiving system including the authentication unit 7008 and thedescramblers 7004 and 7007 will now be described in detail. Thereceiving system receives the scrambled broadcasting contents throughthe tuner 7001 and the demodulating unit 7002. Then, the system manager7015 decides whether the received broadcasting contents have beenscrambled. Herein, the demodulating unit 7002 may be included as ademodulating means according to embodiment of the present invention asdescribed in FIG. 36 to FIG. 65. However, the present invention is notlimited to the examples given in the description set forth herein. Ifthe system manager 7015 decides that the received broadcasting contentshave been scrambled, then the system manager 7015 controls the system tooperate the authentication unit 7008. As described above, theauthentication unit 7008 performs an authentication process in order todecide whether the receiving system according to the present inventioncorresponds to a legitimate host entitled to receive the paidbroadcasting service. Herein, the authentication process may vary inaccordance with the authentication methods.

For example, the authentication unit 7008 may perform the authenticationprocess by comparing an IP address of an IP datagram within the receivedbroadcasting contents with a specific address of a corresponding host.At this point, the specific address of the corresponding receivingsystem (or host) may be a MAC address. More specifically, theauthentication unit 7008 may extract the IP address from thedecapsulated IP datagram, thereby obtaining the receiving systeminformation that is mapped with the IP address. At this point, thereceiving system should be provided, in advance, with information (e.g.,a table format) that can map the IP address and the receiving systeminformation. Accordingly, the authentication unit 7008 performs theauthentication process by determining the conformity between the addressof the corresponding receiving system and the system information of thereceiving system that is mapped with the IP address. In other words, ifthe authentication unit 7008 determines that the two types ofinformation conform to one another, then the authentication unit 7008determines that the receiving system is entitled to receive thecorresponding broadcasting contents.

In another example, standardized identification information is definedin advance by the receiving system and the transmitting system. Then,the identification information of the receiving system requesting thepaid broadcasting service is transmitted by the transmitting system.Thereafter, the receiving system determines whether the receivedidentification information conforms with its own unique identificationnumber, so as to perform the authentication process. More specifically,the transmitting system creates a database for storing theidentification information (or number) of the receiving systemrequesting the paid broadcasting service. Then, if the correspondingbroadcasting contents are scrambled, the transmitting system includesthe identification information in the EMM, which is then transmitted tothe receiving system.

If the corresponding broadcasting contents are scrambled, messages(e.g., entitlement control message (ECM), entitlement management message(EMM)), such as the CAS information, mode information, message positioninformation, that are applied to the scrambling of the broadcastingcontents are transmitted through a corresponding data header or antherdata packet. The ECM may include a control word (CW) used for scramblingthe broadcasting contents. At this point, the control word may beencoded with an authentication key. The EMM may include anauthentication key and entitlement information of the correspondingdata. Herein, the authentication key may be encoded with a receivingsystem-specific distribution key. In other words, assuming that theenhanced data are scrambled by using the control word, and that theauthentication information and the descrambling information aretransmitted from the transmitting system, the transmitting systemencodes the CW with the authentication key and, then, includes theencoded CW in the entitlement control message (ECM), which is thentransmitted to the receiving system. Furthermore, the transmittingsystem includes the authentication key used for encoding the CW and theentitlement to receive data (or services) of the receiving system (i.e.,a standardized serial number of the receiving system that is entitled toreceive the corresponding broadcasting service or data) in theentitlement management message (EMM), which is then transmitted to thereceiving system.

Accordingly, the authentication unit 7008 of the receiving systemextracts the identification information of the receiving system and theidentification information included in the EMM of the broadcastingservice that is being received. Then, the authentication unit 7008determines whether the identification information conform to each other,so as to perform the authentication process. More specifically, if theauthentication unit 7008 determines that the information conform to eachother, then the authentication unit 7008 eventually determines that thereceiving system is entitled to receive the request broadcastingservice.

In yet another example, the authentication unit 7008 of the receivingsystem may be detachably fixed to an external module. In this case, thereceiving system is interfaced with the external module through a commoninterface (CI). In other words, the external module may receive the datascrambled by the receiving system through the common interface, therebyperforming the descrambling process of the received data. Alternatively,the external module may also transmit only the information required forthe descrambling process to the receiving system. The common interfaceis configured on a physical layer and at least one protocol layer.Herein, in consideration of any possible expansion of the protocol layerin a later process, the corresponding protocol layer may be configuredto have at least one layer that can each provide an independentfunction.

The external module may either consist of a memory or card havinginformation on the key used for the scrambling process and otherauthentication information but not including any descrambling function,or consist of a card having the above-mentioned key information andauthentication information and including the descrambling function. Boththe receiving system and the external module should be authenticated inorder to provide the user with the paid broadcasting service provided(or transmitted) from the transmitting system. Therefore, thetransmitting system can only provide the corresponding paid broadcastingservice to the authenticated pair of receiving system and externalmodule.

Additionally, an authentication process should also be performed betweenthe receiving system and the external module through the commoninterface. More specifically, the module may communicate with the systemmanager 7015 included in the receiving system through the commoninterface, thereby authenticating the receiving system. Alternatively,the receiving system may authenticate the module through the commoninterface. Furthermore, during the authentication process, the modulemay extract the unique ID of the receiving system and its own unique IDand transmit the extracted IDs to the transmitting system. Thus, thetransmitting system may use the transmitted ID values as informationdetermining whether to start the requested service or as paymentinformation. Whenever necessary, the system manager 7015 transmits thepayment information to the remote transmitting system through thetelecommunication module 7019.

The authentication unit 7008 authenticates the corresponding receivingsystem and/or the external module. Then, if the authentication processis successfully completed, the authentication unit 7008 certifies thecorresponding receiving system and/or the external module as alegitimate system and/or module entitled to receive the requested paidbroadcasting service. In addition, the authentication unit 7008 may alsoreceive authentication-associated information from a mobiletelecommunications service provider to which the user of the receivingsystem is subscribed, instead of the transmitting system providing therequested broadcasting service. In this case, theauthentication-association information may either be scrambled by thetransmitting system providing the broadcasting service and, then,transmitted to the user through the mobile telecommunications serviceprovider, or be directly scrambled and transmitted by the mobiletelecommunications service provider. Once the authentication process issuccessfully completed by the authentication unit 7008, the receivingsystem may descramble the scrambled broadcasting contents received fromthe transmitting system. At this point, the descrambling process isperformed by the first and second descramblers 7004 and 7007. Herein,the first and second descramblers 7004 and 7007 may be included in aninternal module or an external module of the receiving system.

The receiving system is also provided with a common interface forcommunicating with the external module including the first and seconddescramblers 7004 and 7007, so as to perform the descrambling process.More specifically, the first and second descramblers 7004 and 7007 maybe included in the module or in the receiving system in the form ofhardware, middleware or software. Herein, the descramblers 7004 and 7007may be included in any one of or both of the module and the receivingsystem. If the first and second descramblers 7004 and 7007 are providedinside the receiving system, it is advantageous to have the transmittingsystem (i.e., at least any one of a service provider and a broadcaststation) scramble the corresponding data using the same scramblingmethod.

Alternatively, if the first and second descramblers 7004 and 7007 areprovided in the external module, it is advantageous to have eachtransmitting system scramble the corresponding data using differentscrambling methods. In this case, the receiving system is not requiredto be provided with the descrambling algorithm corresponding to eachtransmitting system. Therefore, the structure and size of receivingsystem may be simplified and more compact. Accordingly, in this case,the external module itself may be able to provide CA functions, whichare uniquely and only provided by each transmitting systems, andfunctions related to each service that is to be provided to the user.The common interface enables the various external modules and the systemmanager 7015, which is included in the receiving system, to communicatewith one another by a single communication method. Furthermore, sincethe receiving system may be operated by being connected with at leastone or more modules providing different services, the receiving systemmay be connected to a plurality of modules and controllers.

In order to maintain successful communication between the receivingsystem and the external module, the common interface protocol includes afunction of periodically checking the status of the oppositecorrespondent. By using this function, the receiving system and theexternal module is capable of managing the status of each oppositecorrespondent. This function also reports the user or the transmittingsystem of any malfunction that may occur in any one of the receivingsystem and the external module and attempts the recovery of themalfunction.

In yet another example, the authentication process may be performedthrough software. More specifically, when a memory card having CASsoftware downloaded, for example, and stored therein in advanced isinserted in the receiving system, the receiving system receives andloads the CAS software from the memory card so as to perform theauthentication process. In this example, the CAS software is read outfrom the memory card and stored in the first memory 7012 of thereceiving system. Thereafter, the CAS software is operated in thereceiving system as an application program. According to an embodimentof the present invention, the CAS software is mounted on (or stored) ina middleware platform and, then executed. A Java middleware will begiven as an example of the middleware included in the present invention.Herein, the CAS software should at least include information requiredfor the authentication process and also information required for thedescrambling process.

Therefore, the authentication unit 7008 performs authenticationprocesses between the transmitting system and the receiving system andalso between the receiving system and the memory card. At this point, asdescribed above, the memory card should be entitled to receive thecorresponding data and should include information on a normal receivingsystem that can be authenticated. For example, information on thereceiving system may include a unique number, such as a standardizedserial number of the corresponding receiving system. Accordingly, theauthentication unit 7008 compares the standardized serial numberincluded in the memory card with the unique information of the receivingsystem, thereby performing the authentication process between thereceiving system and the memory card.

If the CAS software is first executed in the Java middleware base, thenthe authentication between the receiving system and the memory card isperformed. For example, when the unique number of the receiving systemstored in the memory card conforms to the unique number of the receivingsystem read from the system manager 7015, then the memory card isverified and determined to be a normal memory card that may be used inthe receiving system. At this point, the CAS software may either beinstalled in the first memory 7012 upon the shipping of the presentinvention, or be downloaded to the first memory 7012 from thetransmitting system or the module or memory card, as described above.Herein, the descrambling function may be operated by the databroadcasting application manger 7016 as an application program.

Thereafter, the CAS software parses the EMM/ECM packets outputted fromthe demultiplexer 7003, so as to verify whether the receiving system isentitled to receive the corresponding data, thereby obtaining theinformation required for descrambling (i.e., the CW) and providing theobtained CW to the descramblers 7004 and 7007. More specifically, theCAS software operating in the Java middleware platform first reads outthe unique (or serial) number of the receiving system from thecorresponding receiving system and compares it with the unique number ofthe receiving system transmitted through the EMM, thereby verifyingwhether the receiving system is entitled to receive the correspondingdata. Once the receiving entitlement of the receiving system isverified, the corresponding broadcasting service information transmittedto the ECM and the entitlement of receiving the correspondingbroadcasting service are used to verify whether the receiving system isentitled to receive the corresponding broadcasting service. Once thereceiving system is verified to be entitled to receive the correspondingbroadcasting service, the authentication key transmitted to the EMM isused to decode (or decipher) the encoded CW, which is transmitted to theECM, thereby transmitting the decoded CW to the descramblers 7004 and7007. Each of the descramblers 7004 and 7007 uses the CW to descramblethe broadcasting service.

Meanwhile, the CAS software stored in the memory card may be expanded inaccordance with the paid service which the broadcast station is toprovide. Additionally, the CAS software may also include otheradditional information other than the information associated with theauthentication and descrambling. Furthermore, the receiving system maydownload the CAS software from the transmitting system so as to upgrade(or update) the CAS software originally stored in the memory card. Asdescribed above, regardless of the type of broadcast receiving system,as long as an external memory interface is provided, the presentinvention may embody a CAS system that can meet the requirements of alltypes of memory card that may be detachably fixed to the receivingsystem. Thus, the present invention may realize maximum performance ofthe receiving system with minimum fabrication cost, wherein thereceiving system may receive paid broadcasting contents such asbroadcast programs, thereby acknowledging and regarding the variety ofthe receiving system. Moreover, since only the minimum applicationprogram interface is required to be embodied in the embodiment of thepresent invention, the fabrication cost may be minimized, therebyeliminating the manufacturer's dependence on CAS manufacturers.Accordingly, fabrication costs of CAS equipments and management systemsmay also be minimized.

Meanwhile, the descramblers 7004 and 7007 may be included in the moduleeither in the form of hardware or in the form of software. In this case,the scrambled data that being received are descrambled by the module andthen demodulated. Also, if the scrambled data that are being receivedare stored in the third memory 7018, the received data may bedescrambled and then stored, or stored in the memory at the point ofbeing received and then descrambled later on prior to being played (orreproduced). Thereafter, in case scramble/descramble algorithms areprovided in the storage controller 7017, the storage controller 7017scrambles the data that are being received once again and then storesthe re-scrambled data to the third memory 7018.

In yet another example, the descrambled broadcasting contents(transmission of which being restricted) are transmitted through thebroadcasting network. Also, information associated with theauthentication and descrambling of data in order to disable thereceiving restrictions of the corresponding data are transmitted and/orreceived through the telecommunications module 7019. Thus, the receivingsystem is able to perform reciprocal (or two-way) communication. Thereceiving system may either transmit data to the telecommunicationmodule within the transmitting system or be provided with the data fromthe telecommunication module within the transmitting system. Herein, thedata correspond to broadcasting data that are desired to be transmittedto or from the transmitting system, and also unique information (i.e.,identification information) such as a serial number of the receivingsystem or MAC address.

The telecommunication module 7019 included in the receiving systemprovides a protocol required for performing reciprocal (or two-way)communication between the receiving system, which does not support thereciprocal communication function, and the telecommunication moduleincluded in the transmitting system. Furthermore, the receiving systemconfigures a protocol data unit (PDU) using a tag-length-value (TLV)coding method including the data that are to be transmitted and theunique information (or ID information). Herein, the tag field includesindexing of the corresponding PDU. The length field includes the lengthof the value field. And, the value field includes the actual data thatare to be transmitted and the unique number (e.g., identificationnumber) of the receiving system.

The receiving system may configure a platform that is equipped with theJava platform and that is operated after downloading the Javaapplication of the transmitting system to the receiving system throughthe network. In this case, a structure of downloading the PDU includingthe tag field arbitrarily defined by the transmitting system from astorage means included in the receiving system and then transmitting thedownloaded PDU to the telecommunication module 7019 may also beconfigured. Also, the PDU may be configured in the Java application ofthe receiving system and then outputted to the telecommunication module7019. The PDU may also be configured by transmitting the tag value, theactual data that are to be transmitted, the unique information of thecorresponding receiving system from the Java application and byperforming the TLV coding process in the receiving system. Thisstructure is advantageous in that the firmware of the receiving systemis not required to be changed even if the data (or application) desiredby the transmitting system is added.

The telecommunication module within the transmitting system eithertransmits the PDU received from the receiving system through a wirelessdata network or configures the data received through the network into aPDU which is transmitted to the host. At this point, when configuringthe PDU that is to be transmitted to the host, the telecommunicationmodule within the transmitting end may include unique information (e.g.,IP address) of the transmitting system which is located in a remotelocation. Additionally, in receiving and transmitting data through thewireless data network, the receiving system may be provided with acommon interface, and also provided with a WAP, CDMA 1x EV-DO, which canbe connected through a mobile telecommunication base station, such asCDMA and GSM, and also provided with a wireless LAN, mobile internet,WiBro, WiMax, which can be connected through an access point. Theabove-described receiving system corresponds to the system that is notequipped with a telecommunication function. However, a receiving systemequipped with telecommunication function does not require thetelecommunication module 7019.

The broadcasting data being transmitted and received through theabove-described wireless data network may include data required forperforming the function of limiting data reception. Meanwhile, thedemultiplexer 7003 receives either the real-time data outputted from thedemodulating unit 7002 or the data read from the third memory 7018,thereby performing demultiplexing. In this embodiment of the presentinvention, the demultiplexer 7003 performs demultiplexing on theenhanced data packet. Similar process steps have already been describedearlier in the description of the present invention. Therefore, adetailed of the process of demultiplexing the enhanced data will beomitted for simplicity.

The first descrambler 7004 receives the demultiplexed signals from thedemultiplexer 7003 and then descrambles the received signals. At thispoint, the first descrambler 7004 may receive the authentication resultreceived from the authentication unit 7008 and other data required forthe descrambling process, so as to perform the descrambling process. Theaudio decoder 7005 and the video decoder 7006 receive the signalsdescrambled by the first descrambler 7004, which are then decoded andoutputted. Alternatively, if the first descrambler 7004 did not performthe descrambling process, then the audio decoder 7005 and the videodecoder 7006 directly decode and output the received signals. In thiscase, the decoded signals are received and then descrambled by thesecond descrambler 7007 and processed accordingly.

The audio decoder 7005 may receive the signal descrambled by the firstdescrambler 7004 or the signal which is not descrambled by the firstdescrambler 7004 and extract audio data and supplementary informationfrom the audio stream included in the received signal.

The audio decoder 7005 may inversely scale the extracted audio data onthe basis of the scale factor indicated by scale factor indexinformation included in the supplementary information and restore theaudio signal.

The audio decoder 7005 may inversely quantize the audio data, convertthe inversely quantized audio data into a time-domain signal for eachblock according to a block length indicated by an identifier included inthe supplementary information, and restore the audio signal.

The audio decoder 7005 may decode the audio data on the basis of Huffmantable information and region identifying information included in theaudio data and restore the audio signal.

The audio decoder 7005 may receive the signal descrambled by the firstdescrambler 7004 or the signal which is not descrambled by the firstdescrambler 7004 and extract an audio parameter from the audio streamincluded in the received signal. The audio decoder 7005 may calculatesubframe number information included in the audio stream on the basis ofthe extracted audio parameter and calculate start location informationof the subframes on the basis of the calculated subframe numberinformation. The audio decoder 7005 may decode the audio data includedin the subframes according to the audio parameter on the basis of thecalculated start location information.

The audio decoder 7005 may receive the signal descrambled by the firstdescrambler 7004 or the signal which is not descrambled by the firstdescrambler 7004 and extract audio data from the audio stream includedin the received signal. The audio decoder 7005 may inversely quantizethe audio data and restore a stereo audio signal on the basis of middata and side data included in the inversely quantized audio data.

The audio decoder 7005 may receive the signal descrambled by the firstdescrambler 7004 or the signal which is not descrambled by the firstdescrambler 7004, extract a parameter from the audio stream included inthe received signal, decode the extracted parameter, and restore theaudio signal in an extended region. The audio decoder 7005 may decodethe extracted parameter according to algebraic code excited linearprediction (ACELP) and transform coded excitation (TCX) and restore theaudio signal in the extended region. If the parameter includes spectralband replication (SBR), the audio decoder 7005 may decode the parameterand restore the audio signal in a high frequency region of a monochannel.

FIG. 73 is a block diagram showing an MPH receiver according to anembodiment of the present invention. The MPH receiver includes a tuner7300, a demodulator 7310, a demultiplexer 7320, a system information(SI) decoder 7330, a video decoder 7340, and an audio decoder 7350. Theaudio decoder 7350 includes a parser 7351, a core unit 7352 and anextended unit 7353.

The tuner 7300 tunes to the frequency of a specific channel via any oneof an antenna, a cable and a satellite, down-converts the tuned signalinto an intermediate frequency (IF) signal, and outputs the convertedsignal to the demodulator 7310. The received data having the frequencyof the specific channel includes main service data, mobile service dataand table data for decoding the main service data and the mobile servicedata.

In the present embodiment, audio data and video data for a mobilebroadcast may be applied as the mobile service data. Such audio data andvideo data will be compressed by various types of encoders andtransmitted from a broadcasting station. In this case, the audio dataand the video data are decoded by the video and audio decoders 7340 and7350 corresponding to the encoders used for compression such that avideo signal and an audio signal are provided to the user.

The demodulator 7310 performs VSB demodulation and channel equalizationwith respect to the signal output from the tuner 7300, divides thesignal into the main service data and the mobile service data, andoutputs the divided signals in the TS packet units.

The demultiplexer 7320 receives and demultiplexes the data output fromthe demodulator 7310. For example, the demultiplexer 7320 demultiplexesthe mobile service data demodulated by the demodulator 7310 into a datastream, a video stream and an audio stream. Here, the video stream andthe audio stream may be also called a video bitstream and an audiobitstream, respectively. The demultiplexer 7310 demultiplexes systeminformation which is input under the control of the SI decoder 7330. Thesystem information includes mobile service data, a program specificinformation/program and system information protocol (PSI/PSIP) table,and so on. The system information may include channel information, eventinformation and so on.

In the embodiment of the present invention, the PSI/PSIP is applied asthe system information, but the present invention is not limitedthereto. That is, any protocol for transmitting the system informationin a table format is applicable to the present invention regardless ofthe name thereof.

The video and audio decoders 7340 and 7350 may respectively decode thevideo bitstream and the audio bitstream demultiplexed from the mainservice data packet or may respectively decode the video bitstream andthe audio bitstream demultiplexed from the mobile service data packet.According to the embodiment, in the case where the video bitstream andthe audio bitstream as well as the data for the data service areincluded in the mobile service data, the video bitstream and the audiobitstream demultiplexed by the demultiplexer 7320 may be decoded by thevideo decoder 7340 and the audio decoder 7350, respectively.

The parser 7351 of the audio decoder 7350 parses the audio bitstreamoutput from the demultiplexer 7320 and generates audio data and variousparameters necessary for decoding of the audio signal. The audio datamay be time-domain data and or frequency-domain data.

The core unit 7352 is a codec for coding the audio signal excludingsupplementary information. The codec is used even in an encoder forencoding the audio signal as well as a decoder. The core unit may beconfigured in an advanced Audio coding (AAC), MP3, windows media audio(WMA), Oggvorbis or audio coding-3 (AC-3) format and may include a codecwhich will be developed in the future as well as a codec which waspreviously developed if the codec function is performed with respect tothe audio signal. The core unit 7352 processes the audio signal using afilter bank and uses a process such as a block switching process ofchanging the size of the window used for processing the audio signalaccording to the characteristics of the audio signal andquantization/inverse quantization applied to the audio signal. Theprocess performed by the core unit 7352 will be described in detaillater. The core unit 7352 generates the time-domain audio signal usingthe audio data and the parameters output from the demultiplexer 7320.

In addition, for improvement in a bandwidth or improvement in a channel,an extension algorithm may be selectively applied.

The audio decoder 7350 may selectively include the extended unit 7353.The extended unit 7353 indicates a device for processing extendedinformation which is additionally included in the conventional audiosignal format for the improvement in the bandwidth or the improvement inthe channel. The extended information includes the audio data and theparameters for the extension algorithm. For example, extendedinformation for reproducing a multi-channel audio signal may be includedin the supplementary information region of the MPEG-2 or MPEG-4 audioformat. In this case, since there is compatibility with the MPEG-2 orMPEG-4 audio signal, the audio signal including the extended informationmay be used even in the decoder for reproducing only the MPEG-2 orMPEG-4 audio format and the audio signal including the extendedinformation may be used even in the decoder for reproducing themulti-channel audio signal.

The generated time-domain audio signal is output in accordance with asampling frequency and the number of channels. Hereinafter, the reasonwhy the compression of the audio signal is necessary and the basicprinciple used for compressing the audio signal will be described.

FIG. 74 is a view showing a method of compressing an audio signal and anaudio signal processing device for performing the method. FIG. 74A is aschematic view showing a series of processes of encoding the audiosignal by an encoder, transmitting the encoded audio signal, decodingthe encoded audio signal by the decoder, and generating the audiosignal, and FIG. 74B is a view showing the principle of the audio signalcompression performed by the encoder.

Referring to FIG. 74B, an audible frequency of a person is 20 Hz to 20kHz. The audio signal is sampled to a frequency corresponding to twiceor more of a maximum frequency in order to prevent an aliasingphenomenon. A sampling frequency which is generally used is 44.1 kHz or48 kHz. Since an audio signal sample is generally encoded with 16 bits,the sampled audio signal is subjected to the quantization and encodingcoding process and is transmitted at a bitrate of 44.1 kHz * 16 bits=706kbps or 48 kHz * 16 bits=768 kbps.

Since the bitrate is too excessive in the current transmitting systemfor transmitting the audio signal, the bitrate needs to be reduced. Theaudio signal encoder 7400 uses various compression methods in order toreduce the bitrate. If the audio signal is compressed, the bitrate canbe reduced to about 32 to 384 kbps. The compressed encoded audio signalmay be transmitted via a digital channel or stored in a storage medium7410. The audio signal decoder 7420 decodes the compressed encoded audiosignal and outputs the decoded audio signal.

Since a compression method such as a perceptual audio coding (PAC)method is used, the high-quality audio signal can be implemented withlow capacity. In the PAC method, the audio signal is compressed inconsideration of the perceptual capability of the human and a sound ornoise which cannot be heard by the ears of the person is eliminated fromoriginal music or several sounds are synthesized and compressed.

The reason why compression is necessary is as follows: 1) perceptualtransparency can be maintained and the bitrate can be reduced, 2) thebandwidth can be reduced and transmission cost can be reduced, 3)storage requirements can be reduced, and 4) robustness against an errorcan be obtained.

Various PAC methods used for compressing the audio signal are differentfrom one another in sound recognition modeling, sound range filtering,and music data processes according to developers and thus are unlikelyto be compatible with one another. Accordingly, in order to supportvarious types of compression formats, audio decoders should be mountedby the same number of compression formats. Several examples of thevarious compression formats are as follows.

MPEG audio layer 3 (MP3) is a technology which is developed by theMPEG-1 audio standard in the early 1990s and is a digital music formatwhich is most popular among netizens up to now. If the MP3 technology isused, the data amount of audio signal can be compressed to about 1/10 to1/12. The size of the compressed audio file is 2 to 5 MB per music fileand thus high-quality music can be easily exchanged via a network.

The AAC was developed by leaders in audio compression technology usingthe MPEG-2 or MPEG-4 audio standard in 1997 and is excellent incompression performance and quality. Since the size of the audio signalof the AAC format is smaller than that of an MP3 file and the qualitythereof is more excellent than the MP3 file, the AAC is expected to bewidely used as a next-generation compression technology. A maximumsampling frequency is 96 Hz and the number of available channels is 48as a maximum. The compression efficiency of the AAC is about 1.4 timesof that of the MP3.

The AC-3 is a third audio coding method which is developed by DolbyLaboratories, Inc. of the United States of America and is an audioformat having the concept different from the MP3 or AAC. While the MP3or AAC is a compression format based on two channels, the AC-3 is astereophonic support format based on the 5.1 channel (five audiochannels and one low-frequency effect channel). In the AC-3, fivespeakers mounted on the front and back sides, the left and right sides,and the central side and one low-frequency sub woofer speaker are used.The AC-3 is different from the existing analog type surround system inthat the channels are completely separated such that a clean sound canbe delivered without signal interference.

Among methods of reducing the size of digital audio data, there is amethod of reducing the number of bits or a sampling rate as a mostclassical method. However, this method causes a dynamic sound to be lostand generates noise. If the sampling frequency is reduced, the sharpnessof a sound deteriorates. However, if a high-efficiency compressiontechnology is used, the size of the file of the audio signal is reduced,but the amount of digitalized information is not changed.

FIG. 75 is a view illustrating a masking effect used for compressing theaudio signal. The audio encoder encodes the audio signal using themasking effect. The masking effect is one of important characteristicsof sound perception and is a phenomenon that a small sound less than apredetermined threshold is masked by a large sound, that is, aphenomenon that a sound suppresses the perception of another sound. Forexample, the masking effect is a phenomenon that a user cannot hear thevoice of a friend sitting next to the user when a train passes. Themasking effect may be used on the basis of bit allocation when the audiosignal is encoded. For example, the bit may not be allocated to a regionwhich is not perceived due to the masking effect. The masking effectincludes simultaneous masking which is described in the frequency domainand forward masking which is described in the time domain.

The hearing organ of the human analyzes the frequency of the inputsignal by numerous filter banks having different characteristics. Atthis time, the masking effect is generated due to a limitation in theresolution of the hearing organ of the human in the frequency analysisprocess and the signal is perceived by such a preprocessing effect. Theperceived signal is determined by a masking threshold generated by themasking effect.

A masking function is separated and considered according to thecomponents of the audio signal shown in FIG. 80. The masking thresholdis obtained using the masking functions according to the components. Thesound less than the masking threshold is not encoded and quantizationnoise becomes equal to or less than the masking threshold. Hereinafter,the encoder and the decoder for processing the audio signal using themasking threshold will be described.

FIG. 81 is a block diagram showing the basic structure of a generalaudio encoder. The audio encoder includes a time/frequency (T/F) mappingunit 7600, a psycho-acoustic model unit 7610, a bit allocation andquantization unit 7620, and a bitstream packing unit 7630.

The time-domain audio signal is converted into the frequency-domainaudio signal by the T/F mapping unit 7600. At this time, in order togenerate the frequency-domain audio signal in band units, a filter bankmay be used. The psycho-acoustic model unit 7610 eliminates perceptualredundancy using psycho-acoustic modeling, obtains information such asthe masking threshold, and provides bit allocation information used forquantization. The psycho-acoustic modeling unit 7610 converts the audiosignal into the frequency-domain audio signal and computes the maskingthreshold. Next, bits are allocated using a signal-to-masking ratio, asignal-to-noise ratio (SNR) and a noise-to-masking ratio (NMR). Thebitstream packing unit 7630 generates a bitstream using the quantizeddigital audio signal. Although not shown in FIG. 76, the quantized audiosignal is subjected to an entropy coding process before being input tothe bitstream packing unit 7630.

In the entropy coding process, the length of the code representing asymbol varies according to a probability that the symbol is generated.If the code is allocated such that an average code length is closest tothe entropy in the transmission of the digital signal, the efficiency ismost excellent. Accordingly, in the entropy coding process, the averageinformation amount per symbol is decided according to the probabilitythat the symbol is generated, such that the average code length is closeto the entropy. The entropy coding method includes a Huffman codingmethod, an arithmetic coding method and a Lempel-Ziv-Welch (LZW) codingmethod. In the present invention, the audio signal is coded using atleast one of the three entropy coding methods or an entropy codingmethod which will be developed in the future.

The Huffman coding method is one of the entropy coding used fornoiseless compression and is an algorithm using codes having differentlengths according to the appearance frequency of a data character. TheHuffman coding method uses a longer code as the symbol generationprobability is decreased and uses a shorter code as the symbolgeneration probability is increased.

In the Huffman coding method, it is assumed that the probabilities ofthe input symbols are previously known. If the probabilities are notknown, the Huffman coding method may include two steps. In a first step,all the input symbols are read so as to compute the probabilities and,in a second step, the Huffman coding process is performed. The Huffmancoding method has a problem that a probability table should betransmitted together with the compressed data. This is because theencoding apparatus cannot perform the coding process without theprobability table. As a compression rate is increased, the size of theprobability table is increased. In order to solve such problems, anadaptive Huffman algorithm is developed. In the adaptive Huffmanalgorithm, a Huffman tree is adaptively updated while inputting symbols.

In the arithmetic coding method, one codeword is not applied to oneinput symbol, but is applied to all of input symbols. The codewordindicates a value in a range from 0 to 1. That is, a function formapping the input symbols to a value in the range from 0 to 1 isrequired. The arithmetic method is efficient in the case where a binarysymbol is received, the number of input symbols is small, or the symbolgeneration probability is biased.

An adaptive arithmetic coding method is based on a finite-context model.The finite-context model calculates the probability of the input symbolon the basis of the context for generating the symbol. Herein, thecontext indicates a previous input symbol. The case where the previoussymbol is not considered is called order-0, the case where theconsidered previous symbol is 1 is called order-1, the case where theconsidered previous symbol is 2 is called order-2, and the case wherethe considered previous symbol is n is called order-n. However, order-3or more is not generally used. The finite-context model has a problemthat, as the order of the context is linearly increased, a storage spacerequired for storing the context is geometrically increased. That is,the size of storage space used is rapidly increased. Accordingly, thereis a need for efficient management of the storage space.

The LZW coding method is a dictionary-based compression method which isdeveloped for solving the problems of a statistical compression method.In the Huffman coding method, the symbol should be read twice. Incontrast, in the LZW coding method, a code table is made and data iscompressed while the symbol is read once. In the LZW coding method, asmall code table is made in a file and a pattern which is found whilereading the file is added to the table. As a large number of patternsare generated, a large number of code tables are used and a compressionrate is improved. In the LZW coding method, the dictionary indicates astring coded previously. The encoder checks an input string using asliding window. The window is constituted by a search buffer and alook-ahead buffer. The search buffer includes a string coded previouslyand the look-ahead buffer includes a string which will be coded in thefuture. The encoder checks the search buffer while moving a pointer andfinds the location of a string matching with the string included in thelook-ahead buffer. A movement distance of the pointer which is movedfrom the look-ahead buffer is called an offset.

The encoder checks whether the symbol included in the search buffermatches with the symbol included in the look-ahead buffer from a symbolnext to the symbol indicated by the pointer. The length of the stringmatched between the both buffers is called a length of match.

The encoder always finds a longest length of match in the search buffer.If the longest length of match is found, The encoder represents it inthe form of <o, l, c>. Here, o denotes the offset, l denotes the lengthand c denotes the code.

FIG. 77 is a view showing in detail an audio signal encoding apparatusof FIG. 76 according to an embodiment of the present invention.Referring to FIG. 77, the audio signal encoding apparatus includes a T/Fmapping unit 7700, a PA modeling unit 7710, a bit allocation unit 7721,a quantization unit 7722, and a bitstream packing unit 7730. Thedescription of the same components as FIG. 76 will be omitted.

The PA modeling unit 7710 includes a Fast Fourier Transformer (FFT) unit7711 and a masking threshold calculation unit 7712. The audio signal istransformed to the frequency-domain audio signal by the FFT UNIT 7711.Next, the masking threshold calculation unit 7712 calculates a maskingthreshold in the frequency domain using the transformed frequency-domainaudio signal. That is, the perceptual redundancy is eliminated by the PAmodeling unit using the FFT and the masking threshold is then obtainedso as to provide bit allocation information used for quantization.

The FFT is an algorithm for Fourier Transform of discrete data values.The FFT transforms only feature points instead of all of frequencypoints. This is because the feature appears although the feature pointsare extracted and transformed and the remaining portions are compensatedfor.

The bit allocation unit 7721 and the quantization unit 7722 form aniterative loop 7720. That is, a process of performing quantization usingbit values obtained by the psycho-acoustic model is repeated so as todecide an optimal bit number. The bit allocation information is sent tothe bitstream packing unit 7730 as the supplementary information so asto be included in an audio bitstream. The audio signal input to theaudio signal encoding apparatus may be input in the form of a PCM and abitrate is about 768 kbps. In the case where the audio signal is encodedusing the psycho-acoustic model in the audio signal encoding apparatus,the output audio bitstream becomes about 16 to 192 kbps per channel.

FIG. 78 is a block diagram showing an audio signal decoding apparatus.Referring to FIG. 78, the decoding apparatus includes a demultiplexer7800, a lossless decoder 7810 and a synthesis filter bank 7820.

The demultiplexer 7800 parses an audio bitstream transmitted from theencoding apparatus and generates encoded audio signal and supplementaryinformation. The encoded audio signal and supplementary informationexist in each frequency band. Next, the encoded audio signal and thesupplementary information are subjected to a lossless decoding processso as to generate a quantized audio signal. The lossless decodingprocess may use an entropy decoding method. Next, the synthesis filterbank 7820 inversely quantizes and transforms the quantized audio signaland the supplementary information to the time-domain audio signal.

In the process of lossless-decoding and transforming the audio signal tothe time-domain audio signal, the supplementary information may be used.That is, the bit allocation information, the quantization informationand so on may be included in the supplementary information so as to beused in the process of decoding the audio signal. Information forreproducing a multi-channel audio signal, information for reproducing anaudio signal having a three-dimensional (3D) effect, or information forreproducing an audio signal having various ambient effects may beincluded in the supplementary information. Hereinafter, an encodingapparatus using the psycho-acoustic model will be described in detail.

FIG. 79 is a view showing the basic configuration of the encodingapparatus according to the general MPEG standard. Referring to FIG. 79,the encoding apparatus includes a sub-band filter bank 7900, a scalefactor detector 7910, a FFT 7920, a signal-to-masking ratio (SMR)calculator 7930, a bit allocation unit 7940, a quantizer 7950 and abitstream formatter 7960.

The sub-band filter bank 7900 divides a digital audio signal of 168kbps/ch, which is digitalized to 16 bits of 48 kHz, into 32 sub-bands.The scale factor detector 7910 detects scale factors of the 32 sub-bandsof the digital audio signal output from the sub-band filter bank 7900.The FFT unit 7920 Fourier transforms the digital audio signal of 168kbps/ch and outputs the spectrum thereof. The SMR calculator 7930compares the spectrum output from the FFT unit 7920 with the scalefactor detected by the scale factor detector 7910, selects a maximumspectrum of each of the sub-bands, and calculates an SMR using a maskingthreshold and signal power corresponding to the maximum spectrum.

The bit allocation unit 7940 calculates a noise-to-masking ratio (NMR)using the SMR calculated by the SMR calculator and a SNR and allocatesbits according to the NMR. The quantizer 7950 quantizes the digitalaudio signal output from the sub-band filter bank 7900 according to thebits allocated by the bit allocation unit 7940. The bitstream formatter7960 includes the supplementary information in the digital audio signalquantized by the quantizer 7950 and generates a compressed bitstream.The audio signal decoding apparatus reproduces the audio signal by theinverse process of the above-described encoding method.

The FFT unit 7920 and the SMR calculator 7930 use the psycho-acousticmodel. Here, the supplementary information indicates informationnecessary for restoring the compressed quantized digital video signaland includes, for example, scale factor index information and bitallocation information. The scale factor index information may be usedto the bit allocation information according to implementation of theencoder and the decoder. And the scale factor index information mayindicate each of the scale factors.

In more detail, in order to eliminate statistical redundancy, the inputdigital audio signal is converted into sub-band samples by passingthrough the filter bank. The filter bank may be configured by arrangingfilters at the same interval or different intervals according to thefrequency bands as if low frequency regions are densely configured. Theperceptual redundancy is eliminated by the psycho-acoustic model usingthe FFT, the mask threshold is obtained, and bit allocation informationused for quantization is provided.

In layers 1 and 2 of the MPEG, a single sub-band filter bank having 32filters arranged at the same interval is used. Each of the filters usedfor sub-band analysis is based on a 512-tap low-pass filter and thefrequency transitions by a matrix operation such that 32 sub-bandshaving the same size are obtained. The present invention is not limitedto the filter bank and includes filter banks configured by variousmethods.

Generally, the SMR calculated by the psycho-acoustic model may beexpressed by Equation 14 when a ratio of the masking threshold which isthe result of the psycho-acoustic model to the signal power calculatedfrom the scale factor is expressed by dB.

SMR(dB)=signal power(dB)−masking threshold(dB)  Equation 14

where, 32 SMRs corresponding to the sub-bands are obtained in one frame.The physical meaning of the SMR indicates a degree that the signal poweris relatively larger than the masking threshold in each of thesub-bands.

FIG. 86( a) and FIG. 86( b) are a graph showing an SMR curve of thesub-bands in a specific frame according to FIG. 79. As shown in FIG. 80,the SMR has a positive value of 0 dB or more between a sub-band 1 and asub-band 17 and has a negative value of 0 dB or less between a sub-band18 and a sub-band 32. At this time, since all of the signals are alreadymasked in the sub-band section having the negative value of 0 dB or less(for example, between the sub-band 18 and the sub-band 32), the bits areno longer allocated. Accordingly, the bits should be allocated in thesub-band section having the positive value of 0 dB or more (for example,between the sub-band 1 and the sub-band 17).

In order to detect the scale factor for normalizing the sample values ofthe sub-bands, a maximum value of normalized absolute values of 12samples should be found. Next, the maximum value and 64 scale factorssuggested in the MPEG are compared and a scale factor which is largernext to the normalized maximum value is defined as the scale factor ofthe frame.

The bit allocation unit 7940 of FIG. 79 repeatedly performs a process ofallocating one bit to a sub-band having a largest NMR among the 32sub-bands, newly calculating the NMRs of the sub-bands, and allocatingone bit to a sub-band having a largest NMR until a total bit numberallocated to one frame is used up.

The NMR used for the bit allocation process may be expressed by Equation15 using the SNR and the SMR.

NMR(dB)=SMR(dB)−SNR(dB)  Equation 15

where, the SNR denotes a ratio of original signal power to quantizationnoise generated in the quantization process. The physical meaning of theNMR indicates a degree that the quantization noise of a sub-band isrelatively larger than the masking threshold. It can be seen that, asthe NMR is increased, noise to be eliminated is increased.

Accordingly, more bits are allocated to a sub-band having a larger NMRby bit allocation. If one bit is allocated, the SNR is improved by 6 dB.Therefore, the bit allocation indicates a process of allocating the bitsto the sub-bands such that the NMRs thereof become negative values andmaking the quantization noise to be less than the masking threshold.

Accordingly, the noise generated in the encoding process is not causedto be heard by the ears of the person and thus the compression can berealized without the loss of the sound quality.

Since the SMR is a fixed value obtained by the signal power and themasking threshold, Equation 15 is changed by the SNR.

FIG. 81 is a block diagram showing an AAC encoding apparatus accordingto an embodiment of the present invention. Referring to FIG. 81, the AACencoding apparatus includes a psycho-acoustic modeling unit 8100, apre-processing unit 8110, a filter bank 8120, a temporal noise shaping(TNS) unit 8130, an intensity/coupling unit 8140, a prediction unit8150, a mid/side (M/S) stereo processing unit 8160, a data restorationunit 8170 and a bitstream formatter 8180. The data restoration unit 8170has an iteration loop structure and includes a rate/distortion controlunit 8171, a scale factor extractor 8172, a quantizer 8173 and anoiseless coding unit 8174.

Pulse code modulation (PCM) audio signals are input to thepsycho-acoustic modeling unit 8100 and the pre-processing unit 8110. Thepre-processing unit 8110 changes the sampling frequencies of the inputaudio signals according to the bitrate and outputs the audio signals. Insome cases, the pre-processing unit 8110 may not be included.

The psycho-acoustic modeling unit 8100 collects signals having properscale factors of the input audio signals and calculates the maskingthresholds of scale factor bands using a masking phenomenon generated byinteraction of the signals. The outputs of the psycho-acoustic modelingunit 8100 are input to the filter bank 8120, the TNS unit 8130, theintensity/coupling unit 8140 and the M/S stereo processing unit 8160.

The filter bank 8120 eliminates the statistical redundancy of the audiosignal using the masking threshold of the psycho-acoustic modeling unit8100 and the signal output from the pre-processing unit 8110 and outputsthe signal to the TNS unit 8130. That is, the filter bank 8120 dividesthe overall band of the audio signal into 32 sub-bands having the samefrequency width and codes the 32 sub-bands to sub-band samples. In thepresent invention, the filter bank for dividing the overall frequencyinto the sub-bands having different intervals or the filter bank fordividing the overall frequency into the sub-bands having the sameinterval may be used.

The sub-band samples coded by the filter bank 8120 are input to the datarestoration unit 8170 via the TNS unit 8130, the intensity/coupling unit8140, the prediction unit 8150 and the M/S stereo processing unit 8160.The TNS unit 8130, the intensity/coupling unit 8140, the prediction unit8150 and the M/S stereo processing unit 8160 may be selectively used inthe encoder.

That is, the TNS unit 8130 receives the output of the filter bank 8120and the psycho-acoustic modeling unit and controls the temporal shape ofthe quantization noise in each window used for transformation. At thistime, the temporal noise shaping can be realized by applying a processof filtering frequency data.

The intensity/coupling unit 8140 is a module for more efficientlyprocessing a stereo signal, which receives the output of thepsycho-acoustic modeling unit 8100 and the output of the TNS unit 8130,codes only quantized information of the scale factor band of one of twochannels and transmits only the scale factor of the other channel. Theintensity/coupling unit 8140 is not necessarily used in the encoder, andit is checked whether or not the intensity/coupling unit is used in theunits of the scale factor band, in consideration of various matters.

The prediction unit 8150 receives the output of the intensity/couplingunit 8140 and the output of the quantizer 8173 of the data restorationunit 8170 and predicts frequency coefficient values of a current frame.A difference between the predicted value and an actual frequencycomponent is quantized and coded such that the amount of generated bitscan be reduced. At this time, the prediction unit 8150 may beselectively used in frame units. That is, since complexity forpredicting a next frequency coefficient is increased if the predictionunit 8150 is used, the prediction unit may not be used. In some cases, aprediction difference may be larger than an original signal and thus theamount of bits which are actually generated by the prediction may belarger compared with the case where the prediction is not performed. Atthis time, the prediction unit 8150 is not used.

The M/S stereo processing unit 8160 is a module for more efficientlyprocessing the stereo signal, which receives the output of thepsycho-acoustic modeling unit 8100 and the output of the prediction unit8150, transforms the outputs to a sum signal of or a difference signalbetween a left channel signal and a right channel signal, and processesthe signals. The M/S stereo processing unit 8160 is not necessarily usedin the encoder and it is checked whether or not the M/S stereoprocessing unit is used in the units of the scale factor band, inconsideration of various matters. The output of the M/S stereoprocessing unit 8160 is input to the scale factor extractor 8172 of thedata restoration unit 8110.

The scale factor extractor 8172 extracts scale factors under the controlof the rate/distortion control unit 8171, normalizes the sub-bandsamples output from the M/S stereo processing unit 8160, and outputs thenormalized sub-band samples to the quantizer 8173.

The quantizer 8173 quantizes the sub-band samples normalized by thescale factor extractor 8172 and outputs the quantized sub-band samplesto the noiseless coding unit 8174. That is, the quantizer 8173 quantizesthe sub-band samples of each band such that the level of thequantization noise of each band becomes less than the masking thresholdso as not to be heard by the human.

The signal quantized by the quantizer 8173 is coded by the noiselesscoding unit 8174 and is output to the rate/distortion control unit 8171and the bitstream formatter 8180. The bitstream formatter 8180 collectsinformation of the modules 8110 to 8170, and forms and transmits abitstream.

In the AAC coding process of FIG. 81, the structure for obtaining themasking threshold in the frequency domain by the psycho-acoustic modeland allocating the bits on the basis of the masking threshold such thatthe quantization noise is not heard is equal to that of the MP3 codingprocess. However, in the AAC coding process, the frequency convertingprocess is different from that of the MP3 in that a 2048-point modifieddiscrete cosine transform (MDCT) having a higher frequency resolutionthan that of the MP3 is performed (filter bank), the TNS which is atechnology of shaping the noise in the time domain is used, and theprediction for eliminating the statistical redundancy between the framesis used. In the AAC coding process, a pre-processing unit 8110 forchanging the sampling frequency according to the transmission bitrate isincluded.

In such a structure, the noiseless coding unit 8174 of the AAC uses aHuffman coding method. The Huffman coding method is similar to the MP3in that two or four MDCT coefficients (n-tuples) are coded using aHuffman codebook, but is different from the MP3 in that a region usingthe same codebook is divided into sections and a rzero region and acount1 region do not exist. That is, in the AAC coding process, thesections are divided so as to bind the regions having similarstatistical characteristics with respect to the MDCT spectrumcoefficient and most suitable codebooks of the sections are selected soas to perform the Huffman coding process. At this time, the number ofHuffman codebooks used for the AAC is total 12.

In the AAC, a maximum value which can be expressed by a Huffman table is15 and an escape coding (ESC) process should be performed in order toexpress a value larger than the maximum value. That is, in order to codeinformation of 16 or more, a codeword corresponding to 16 is Huffmancoded and then escape_sequence of Equation 16 is prefixed.

escape_sequence=<escape_prefix><escape_separator><escape_word>  Equation16

where, <escape_prefix> denotes an N-bit binary number “1”,<escape_separator> denotes a binary number “0”, and <escape_word>denotes a coefficient expressed by an unsigned integer. Accordingly, Nis a minimum value which can express the MDCT coefficient in which2(N+4) is located at <escape_word>.

While the number of bits used for performing the coding process by theESC process is decided by selecting the Huffman codebook in the MP3, thenumber of bits used is decided according to the coefficients of the ESCprocess in the AAC. Hereinafter, the AAC decoding apparatus will bedescribed.

FIG. 82 is a block diagram showing an AAC decoding apparatus accordingto an embodiment of the present invention. Referring to FIG. 82, the AACdecoding apparatus includes a bitstream formatter 8200, a decoding andinverse quantization unit 8210, an M/S stereo processing unit 8220, aprediction unit 8230, an intensity/coupling unit 8240, a TNS unit 8250,a normalization unit 8260, a filter bank 8270 and an AAC gain controltool 8280.

If a coded audio bitstream is received, the bitstream formatter 8200extracts information necessary for decoding by a demultiplexing processand the decoding and inverse quantization unit 8210 performs the Huffmandecoding and inverse quantization process. Next, the audio signal isreproduced by the M/S stereo restoration, prediction, intensity/couplingstereo restoration, TNS filtering, normalization, filter bank and AACgain control process. The decoding process is equal to or similar to thecoding process of FIG. 81 and thus the detailed description thereof willbe omitted.

The M/S stereo processing unit 8220, the prediction unit 8230, theintensity/coupling unit 8240, the TNS unit 8250 and the normalizationunit 8260 process the frequency-domain audio signal and the audio signaloutput from the filter bank 8270 is transformed to the time-domain audiosignal. Hereinafter, the Huffman coding process will be described.

FIG. 83 is a flowchart illustrating a double structure for bitallocation and quantization according to an embodiment of the presentinvention. Referring to FIG. 83, the process of performing thequantization and Huffman coding by bit allocation in the iterative loopwill be described. The iterative loop includes a double structure(called a nested loop) of a distortion control loop (external loop)(step 8310) of comparing the quantization noise with the maskingthreshold and performing noise shaping and a quantization loop (internalloop) (step 8300) of comparing a bit allocation result for thequantization (step 8301) and coding (step 8302) with the number ofavailable bits. The MDCT coefficient to be quantized, the maskingthreshold which is the output of the psycho-acoustic model and thenumber of available bits decided by the compression bitrate are input tothe iterative loop. In the quantization loop of the iterative loop,nonlinear quantization of the MDCT coefficient is performed.

The MDCT coefficient quantized by the nonlinear quantization process(step 8301) is Huffman coded (step 8302). By this step, the frame iscoded and the number of bits used is calculated. If the number of bitsused is larger than the number of available bits, it is indicated thatthe bits are excessively used. Thus, the quantization step size isadjusted and the quantization and Huffman coding process is performed.If the condition is satisfied, the process escapes from the internalloop (step 8300) (step 8303). If the quantization and Huffman codingprocess is performed by the internal quantization loop, the inversequantization from the quantized MDCT coefficient is performed and adistortion, that is, a quantization noise value, is calculated from adifference between the inversely quantized MDCT coefficient and theoriginal real MDCT coefficient in the external distortion control loop.The calculated distortion is compared with the masking threshold whichis the output of the psycho-acoustic model in the frequency domain unitsuch as a scale factor band (step 8330). If the distortion value islarger than the masking threshold (that is, the distortion is heard bythe ears of the person), the scale factor value of the band is updated(step 8320) and the process returns to the internal quantization loop.In the internal loop, the iterative loop for performing the nonlinearquantization and Huffman coding process is performed with respect to theDMCT coefficient xr(n) which is newly decided by the scale factor. Ifthe condition of the masking threshold and the distortion of all of thebands are satisfied, the iterative loop is completed. Since theiterative loop is repeated until the escaping condition of the externalloop and the escaping condition of the internal loop are satisfied, theiterative loop for the bit allocation and the quantization has a largeoperation amount.

The quantization and Huffman coding process should be performed in theinternal quantization loop due to the Huffman coding. Accordingly, sincethe number of bits used actually is not decided by factors such as aquantization step and can be checked after the quantization and Huffmancoding process is performed, the quantization and Huffman coding processis repeated until an optimal value is obtained. Accordingly, a largeoperation amount is required for the performing of the process.

FIG. 84 is a graph showing a bit allocation method according to anembodiment of the present invention. Referring to an upper graph of FIG.84, a graph shown by a thin solid line denotes an original audio signaland the upper graph shown by a thick solid line denotes the maskingthreshold.

A lower graph of FIG. 84 shows the ratio (SMR) of the power of the audiosignal shown in the upper graph to the power of the masking threshold.FIG. 84 shows the SMR of each band. That is, in FIG. 84, the frequencydomain is divided into 32 bands and the SMR value of each of the bandsis shown. The SMRs of the bands located in the left region have positivevalues having small levels, the SMRs of the bands located in the middleregion have positive values having large levels, and the SMRs of thebands located in the right region have negative values having largelevels. The region in which the SMRs have the negative values does notneed to be transmitted because the sound is not heard due to the maskingeffect. A small number of bits is allocated and coded in the region inwhich the SMRs have the positive values having the small levels and alarge number of bits is allocated and coded in the region in which theSMRs have the positive values having the large levels.

The masking threshold may be calculated using the psycho-acoustic modelof the audio signal and the bit value necessary for coding the audiosignal using the masking threshold may be allocated. In particular,since the portion in which the SMRs have the negative value does notneed to be transmitted, the data amount of the audio signal to betransmitted can be significantly reduced.

Accordingly, in the present invention, the encoding process causes thebands of the audio signal to have different quantization noises byweighting based on the masking threshold. That is, different bits can beallocated to the bands of the audio signal. For example, in the casewhere the AAC and the MP3 are used, the scale factor can be scaledaccording to the scale factor bands on the basis of the maskingthreshold.

In the decoding process, the bit allocation information generated in thecoding process is received and the spectrum of the audio signal isinversely quantized on the basis of the bit allocation information.Herein, the scale factor index information may be used to the bitallocation information. And the scale factor index information mayindicate each of the scale factors.

For example, in the case where the AAC and the MP3 are used, the levelof the audio signal can be restored by the scale factor transmittedaccording to the scale factor bands. That is, the audio signal can bedecoded using the generated scale factor such that the quantizationnoise varies according to the bands.

The reason why the MP3 and AAC coding process is more complicated thanthe decoding process is because the active bit allocation using thepsycho-acoustic model has a large operation amount. The psycho-acousticmodel includes a large amount of exponential or log operations which isinefficient for implementing a digital signal processor (DSP) and thebit allocation process requires much time because the nonlinearquantization and Huffman coding process is repeatedly performed severaltimes until optimal noise shaping is realized. The operation amountoccupied by the bit allocation and the psycho-acoustic model operationin the MP3 and AAC is as large as 85% of the total operation amount. Inparticular, since the operation amount is significantly changedaccording to the characteristics of the input data in the bit allocationprocess, there is a difficulty in a real-time process of the audiosignal. Hereinafter, the nonlinear quantization included in theiterative loop will be described.

The quantization indicates approximation of a value in a real-numberregion to specific allowable values. In the quantization process, anerror corresponding to the difference between the original signal andthe quantized signal occurs and is called a quantization error. Thequantization error can be reduced by reducing the quantization stepsize. A constant quantization step size is called a linear quantizationand a variable quantization step size is called a nonlinearquantization. In the linear quantization, the level of the error isconstant regardless of the level of the input signal. Since a relativeerror is reasonably uniform, a nonlinear quantization method for varyingthe quantization step size is advantageous. For example, thequantization step size is varied such that the quantization step size isdecreased with respect to a signal having a small amplitude and isincreased with respect to a signal having a large amplitude.

Equation for nonlinear quantization is expressed by Equation 17.

$\begin{matrix}{{{is}(n)} = \left\lbrack {\left( \left( \frac{{xr}(n)}{quant} \right)^{({3/4})} \right) - 0.0946} \right\rbrack} & {{Equation}\mspace{14mu} 17}\end{matrix}$

where, n denotes an MDCT coefficient index of 0, 1, 2, . . . , and N−1and N denotes 556 in the MP3 and 1024 in the AAC. xr(n) denotes an MDCTcoefficient having a real number value and is(n) denotes a quantizedMDCT coefficient. In Equation 17, quant is called the quantization stepsize or a global gain and is a value which is a control target value inthe quantization loop because the bit number and the quantization noiseused for coding the frame is decided by the actual quant value.

As described above, the reason why the nonlinear quantization isperformed is because a signal having a low level is more preciselyquantized than a signal having a high level, that is, because thequantization error varies according to the level of the audio signal. Inthe case where the nonlinear quantization is used, as expressed byEquation 17, the signal having the high level is reduced by a power of3/4. In the decoding apparatus, the xr value is obtained by a power of4/3.

Generally, in the audio signal, the signal having the low level isfrequently generated and the signal having the high level is notfrequently generated. Accordingly, it is advantageous that more bits areused in the signal having the low level, compared with the case wherethe same number of bits is used in the signal having the low level andthe signal having the high signal. Accordingly, the encoding apparatusaccording to the present invention uses the nonlinear quantization andthe decoding apparatus according to the present invention parses theaudio bitstream and inversely quantizes the quantized value of eachspectrum. At this time, the audio signal can be restored by an inverseexponential operation using a power of 4/3 as described above.Hereinafter, the MP3 encoding apparatus using the bit allocation and thenonlinear quantization will be described.

FIG. 85 is a block diagram showing an MP3 encoding apparatus accordingto an embodiment of the present invention.

Referring to FIG. 85, the MP3 encoding apparatus includes a sub-bandfilter bank 8510, the MDCT unit 8520, a distortion control unit 8550, aFFT unit 8530, a psycho-acoustic model unit 8540 and a bitstreamformatter 8560.

The distortion control unit 8550 is an iterative loop and includes anonlinear quantizer 8552, a bit allocation unit 8551, a Huffman codingunit 8554, and a supplementary information coding unit 8553.

First, the audio signal is input to the sub-band filter bank 8510 inorder to eliminate the statistical redundancy and, at the same time, isinput to the FFT unit 8530 in order to eliminate perceptual redundancydue to the hearing characteristics of the human.

The sub-band filter bank 8510 may be constituted by 32 filter bankswhich are arranged at the same interval by a weighting overlappingadding method. The audio signal is transformed to sub-band samples bythe sub-band filter bank 8510. The sub-band samples of which thestatistical redundancy of the audio signal is eliminated by the sub-bandfilter bank 851 are output to the MDCT unit 8520, the MDCT is performed,and a hybrid transform coding method for increasing the frequencyresolution is used.

That is, in order to efficiently use the hearing characteristics, thesignal is divided by the frequency component. Accordingly, the sub-bandfilter bank 8510 codes the audio signal of which the overall frequencyis divided into 32 sub-bands having the same-interval frequency width.

The FFT unit 8530 transforms the input audio signal into thefrequency-domain audio signal via the FFT and outputs the transformedaudio signal to the psycho-acoustic model unit 8540. The psycho-acousticmodel unit 8540 using the FFT obtains the masking threshold which is thenoise level from the FFT frequency signal in order to eliminate theperceptual redundancy due to the hearing characteristics of the humanfrom the input audio signal.

As described above, the masking phenomenon is one of the importantcharacteristics of sound perception and is a phenomenon that a smallsound having a predetermined threshold or less is masked by a largesound, that is, a phenomenon that one sound is suppressed from beingheard by another sound. In such a masking phenomenon, a frequencymasking phenomenon appears when two sounds are simultaneously generated.That is, the frequency masking causes a sound, which is masked when asound having a specific frequency masks a sound having a differentfrequency, to be heard when the masked sound has energy of apredetermined threshold or more. The threshold used at that time is themasking threshold and is different from an absolute threshold. Theabsolute threshold indicates a threshold which can perceive any sound.

The bit allocation unit 8551 of the distortion control unit 8550allocates the quantization bits of the MDCT coefficient on the basis ofthe masking threshold of each frame and outputs the bits to thequantizer 8552. At this time, a minimum number of bits are allocated tothe sub-band samples such that the quantization noise is masked by theaudio signal.

The bit allocation, the quantization and the Huffman coding process ofthe distortion control unit 8550 is performed by the iterative loop. Atthis time, the bit allocation unit 8551 allocates the bits according tothe scale factor bands such that the subjective noise is minimized whenthe noise cannot be completely masked. The MDCT coefficient which isnonlinearly quantized by the quantizer 8552 on the basis of theallocated bits is input to the Huffman coding unit 8554, which issubjected to the Huffman coding process and is output to the bitstreamformatter 8560. The Huffman coded audio signal is transformed to thebitstream by the bitstream formatter 8560 together with thesupplementary information and the bitstream is transmitted.

Here, the Huffman coding method is one of the statistical compressionmethods, which allocates a bit number representing unit information onthe basis of appearance frequency of the unit information. That is,information having a high appearance frequency is represented by a smallnumber of bits and information having a low appearance frequency isrepresented by a large number of bits, such that the number of bitsnecessary for representing the overall data is reduced. Accordingly, avariable-length code of which the length varies according to the unitinformation is used for audio and video compression. Since the datacompressed by the Huffman coding method can restore original data by theHuffman coding method without an error, the Huffman coding method is anoiseless coding method.

In the MP3 and AAC coding process, a method of performing the Huffmancoding process with respect to the quantized signal, finding a codebookindex corresponding to the unit information using a Huffman codebook,and transmitting a codeword corresponding thereto is used. The audiosignal may be coded by a joint stereo method in order to reduce abitrate. Hereinafter, the joint stereo method will be described.

FIG. 86 is a view showing equations for illustrating the joint stereomethod according to an embodiment of the present invention.

Generally, the stereo signal method indicates a method of coding theaudio signal to two independent channels and reproducing the audiosignal using left and right speakers. That is, the stereo signalchannels are completely independent data. In contrast, the joint stereomethod indicates a method for eliminating the correlation betweenchannels, that is, a method of sharing the common portion of the leftand right channels so as to reduce the amount of data and coding theaudio signal. The amount of data transmitted is reduced due to sharing.The reduced data is replaced with data used for improving sound quality,such that the sound quality can be further improved than the stereomethod if the same band is used.

For example, in 128-kbps stereo, one channel becomes 64 kbps. In thejoint stereo, if it is assumed that 20 Kbps is shared, 10 Kbps perchannel can be saved. Accordingly, if the joint stereo is used, theaudio signal can be coded so as to have the same sound quality as thestereo channel coded by 79 Kbps per channel. The joint stereo methodincludes an intensity stereo method and a mid/side (M/S) stereo method.

FIG. 86A shows an equation for expressing the intensity stereo method.In the intensity stereo method, only the spectrum information of one ofthe left and right channels is coded and transmitted and a power ratioof the both channels is transmitted. The other channel copies thespectrum information of the transmitted channel and restores thespectrum information using the power ratio. For example, as shown inFIG. 86A, only the spectrum information of the left channel istransmitted. At this time, the spectrum information of the left channelis generated by adding the spectrum information of the right channel andthe spectrum information of the left channel (L_(i)=R_(i)+L_(i)). Thespectrum information of the right channel is not transmitted (R_(i)=0).Instead, the ratio of the power of the left channel to the power of theright channel is transmitted. The power information is associated withenvelope information of the audio signal and the envelope informationcan be obtained by the scale factor.

The spatial position of the sound source is obtained by a powerdifference in a high frequency signal and is obtained by a phasedifference in a low frequency signal. Accordingly, the intensity stereomethod may be used in the high frequency region.

The encoding apparatus can generate information on the frequency band towhich the intensity stereo method is applied and flag informationindicating whether or not the intensity stereo method is used accordingto the frames or bands. The encoding apparatus can reproduce the audiosignal using the information on the frequency band to which theintensity stereo method is applied and the flag information indicatingwhether or not the intensity stereo method is used according to theframes or bands.

FIG. 86B shows an equation showing the M/S stereo coding method. In theM/S stereo method, the audio signal is coded using the sum of and thedifference between two channels. For example, as shown in FIG. 86B, theright channel signal and the left channel signal are added to generate amid signal M_(i) and the left channel signal is subtracted from theright channel signal to generate a side signal S_(i). The encodingapparatus codes the generated mid signal and side signal and transmitsthe coded signals, and the decoding apparatus restore the original rightchannel signal and left channel signal using the transmitted mid signaland side signal.

Like the intensity stereo method, the encoding apparatus can generateinformation on the frequency band to which the M/S stereo method isapplied and flag information indicating whether or not the M/S stereomethod is used according to the frames or bands. The decoding apparatuscan reproduce the audio signal using the information on the frequencyband to which the M/S stereo method is applied and the flag informationindicating whether or not the M/S stereo method is used according to theframes or bands. The intensity stereo method and the M/S stereo methodare selectively used and cannot be simultaneously with respect to thesame signal.

In another embodiment of the transmitting system, mobile service data istransmitted on the basis of an Internet protocol (IP).

In one embodiment of the present invention, main service data istransmitted on the basis of an MPEG-2 and mobile service data istransmitted on the basis of the IP.

FIG. 87 shows another embodiment of a protocol stack for providing theIP-based mobile service in order to prevent the above-described problem.

In FIG. 87, an adaptation layer is interposed between the IP layer andthe physical layer such that the IP datagram and the PSI/PSIP data aretransmitted without using the MPEG-2 TS format.

Even in FIG. 87, the signaling for the mobile service is encapsulated tothe PSI/PSIP section structure and is referred to as PSI/PSIP data, forconvenience of description. A RTP header, a UDP header, and an IP headerare sequentially prefixed to the A/V payload for the mobile service soas to configure the IP datagram as shown in FIG. 88. That is, Audiostream may be included in A/V Payload field of the IP datagram as shownin FIG. 88 and may be transmitted to a broadcast receiver. And thebroadcast receiver receives the IP datagram including the Audio streamand restores audio signal by extracting an audio stream from the IPdatagram.

The adaptation layer is a data link layer for dividing the IP datagramand the PSI/PSIP data of the section type and linking the data such thatthe divided data can be processed in the upper layer.

That is, in the adaptation layer, a RS frame including the PSI/PSIPdata, the IP datagram and an identifier for identifying the PSI/PSIPdata and the IP datagram is generated.

As described above, the present invention has the following advantages.More specifically, the present invention is highly protected against (orresistant to) any error that may occur when transmitting supplementaldata through a channel. And, the present invention is also highlycompatible to the conventional receiving system. Moreover, the presentinvention may also receive the supplemental data without any error evenin channels having severe ghost effect and noise.

Furthermore, the present invention is even more effective when appliedto mobile and portable receivers, which are also liable to a frequentchange in channel and which require protection (or resistance) againstintense noise.

It will be apparent to those skilled in the art that variousmodifications and variations can be made in the present inventionwithout departing from the spirit or scope of the inventions. Thus, itis intended that the present invention covers the modifications andvariations of this invention provided they come within the scope of theappended claims and their equivalents.

1. A broadcasting receiver comprising: a receiver which receives abroadcasting signal including main service data and mobile service data,wherein the mobile service data configures a data group, and the datagroup is divided into a plurality of regions, known data streams arelinearly inserted into some of the plurality of regions, andinitialization data used for initializing a memory in a trellis encoderof a transmitter is located at a start portion of the known data stream;a known data detector which detects known data in the receivedbroadcasting signal; an equalizer which channel-equalizes the receivedmobile service data using the detected known data; and a decoder whichextracts audio data and supplementary information from thechannel-equalized mobile service data, inversely scales the audio dataon the basis of a scale factor indicated by scale factor indexinformation included in the supplementary information, and restores anaudio signal.
 2. The broadcasting receiver according to claim 1, whereinN known data streams are inserted into some of the plurality of regionsand a transfer parameter is inserted between a first known data streamand a second known data stream among the N known data streams.
 3. Thebroadcasting receiver according to claim 1, wherein the mobile servicedata configures a RS frame, and the RS frame includes at least one datapacket of the mobile service data, a RS parity generated based on the atleast one data packet, and a CRC checksum generated based on the atleast one data packet and the RS parity.
 4. The broadcasting receiveraccording to claim 1, wherein the audio data is nonlinearly quantized.5. The broadcasting receiver according to claim 4, wherein the decoderperforms a reciprocal exponentiation and inversely quantizes the audiodata.
 6. The broadcasting receiver according to claim 5, wherein thedecoder performs the reciprocal exponentiation using 4/3 as a reciprocalexponent.
 7. The broadcasting receiver according to claim 1, wherein thedecoder inversely scales the audio data on the basis of the scale factorallocated to bands of the audio data.
 8. The broadcasting receiveraccording to claim 1, wherein the decoder includes: a bitstreamformatter which extracts the audio data and the supplementaryinformation from the channel-equalized mobile service data; a decodingand inverse quantization unit which inversely scales the audio data onthe basis of the scale factor indicated by the scale factor indexinformation included in the supplementary information; and anintensity/coupling unit which calculates stereo audio data on the basisof information on a ratio of the power of the inversely scaled audiodata to the power included in the supplementary information.
 9. A methodof processing a broadcasting signal, the method comprising: receiving abroadcasting signal including main service data and mobile service data;detecting known data in the received broadcasting signal;channel-equalizing the received mobile service data using the detectedknown data; and extracting audio data and supplementary information fromthe channel-equalized mobile service data, inversely scaling the audiodata on the basis of a scale factor indicated by scale factor indexinformation included in the supplementary information, and restoring anaudio signal, wherein the mobile service data configures a data group,and the data group is divided into a plurality of regions, known datastreams are linearly inserted into some of the plurality of regions, andinitialization data used for initializing a memory in a trellis encoderof a transmitter is located at a start portion of the known datastreams.
 10. The method according to claim 9, wherein N known datastreams are inserted into some of the plurality of regions and atransfer parameter is inserted between a first known data stream and asecond known data stream among the N known data streams.
 11. The methodaccording to claim 9, wherein the mobile service data configures a RSframe, and the RS frame includes at least one data packet of the mobileservice data, a RS parity generated based on the at least one datapacket, and a CRC checksum generated based on the at least one datapacket and the RS parity.
 12. The method according to claim 9, whereinthe audio data is nonlinearly quantized.
 13. The method according toclaim 12, wherein the decoding step includes performing a reciprocalexponentiation and inversely quantizes the audio data.
 14. The methodaccording to claim 9, wherein the decoding step includes inverselyscaling the audio data on the basis of the scale factor allocated tobands of the audio data.